System and method for demodulation of resolver outputs

ABSTRACT

Demodulation circuitry includes an input terminal configured to be coupled to an analog-to-digital converter (ADC) and configured to receive a plurality of ADC outputs. The plurality of ADC outputs are generated based on resolver outputs. The demodulation circuitry also includes a rectifier configured to rectify the plurality of ADC outputs. Rectifying the plurality of ADC outputs preserves a phase of the plurality of ADC outputs. The demodulation circuitry includes amplitude determination circuitry configured to determine, based on the rectified plurality of ADC outputs, demodulated amplitude values corresponding to the resolver outputs. The demodulation circuitry further includes angle computation circuitry configured to generate position outputs based on the demodulated amplitude values.

FIELD OF THE DISCLOSURE

The present disclosure is generally related to demodulation of resolveroutputs and gimbaled inertial measurement units.

BACKGROUND

A gimbaled inertial measurement unit is used for vehicle navigation andobject tracking. The gimbaled inertial measurement unit includesmultiple gimbals that each rotate along a single axis to positionsensors along the vehicle's path. By using multiple gimbals, such asthree or four gimbals, a vehicle's inertia can be monitored in multipleaxes by the sensors and used for inertial navigation. Inertialnavigation continuously calculates by dead reckoning the position, theorientation, and the velocity of a moving object without the need forexternal reference. Dead reckoning (or deductive reckoning) involvescalculating the vehicle's current position by using a previouslydetermined position and advancing that position based upon estimatedspeeds and headings.

Gimbaled inertial measurement units include additional sensors, such asresolvers, to determine a position of a motor that drives each of themultiple gimbals and positions the sensors that track the vehicle'sinertia. A resolver is an analog sensor that is used to determinerotational position, such as an angle. The resolver receives anexcitation signal and generates analog output signals which areconverted to digital samples. The digital samples are used to determinea position of the motor and the gimbal. The position outputs determinedfrom the output analog signals may lose precision during the conversionfrom analog to digital or from processing the digital samples into theangle outputs. Additionally, interference from current switching duringoperating the gimbal motors can add noise and errors. Reduced precisionand errors accumulate over time in inertial navigation and can lead toincorrect data or navigation.

Complicated discrete solutions are often used to increase precision orbandwidth of the gimbaled inertial measurement units needed for missionor design requirements. However, these solutions do not generallyprovide increased precision and increased bandwidth simultaneously.Additionally, these solutions add complexity, cost, weight, and volumeto vehicle design. In the context of flying vehicles (aircraft,spacecraft, etc.), weight and volume greatly increase cost and reduceperformance.

SUMMARY

In a particular implementation, an apparatus includes a coarse resolverconfigured to output coarse position signals indicative of a coarseposition of a drive shaft of a motor. The apparatus also includes a fineresolver configured to output fine position signals indicative of a fineposition of the drive shaft of the motor. The apparatus further includesa control circuit. The control circuit is configured to receive thecoarse position signals from the coarse resolver and the fine positionsignals from the fine resolver and generate an initial position output,based on the coarse position signals, that indicates an initial positionof the drive shaft. The control circuit is further configured togenerate a subsequent position output, based on the fine positionsignals, that indicates a subsequent position of the drive shaft.

In another particular implementation, a method of determining rotationalposition includes receiving coarse position signals from a coarseresolver and fine position signals from a fine resolver. The coarseposition signals are indicative of a coarse position of a drive shaft ofa motor, and the fine position signals are indicative of a fine positionof the drive shaft of the motor. The method also includes generating aninitial position output, based on the coarse position signals, thatindicates an initial position of the drive shaft. The method furtherincludes generating a subsequent position output, based on the fineposition signals, that indicates a subsequent position of the driveshaft.

In yet another particular implementation, a non-transitory computerreadable medium stores instructions that, when executed by a processor,cause the processor to receive coarse position signals from a coarseresolver and fine position signals from a fine resolver. The coarseposition signals are indicative of a coarse position of a drive shaft ofa motor, and the fine position signals are indicative of a fine positionof the drive shaft of the motor. The instructions further cause theprocessor to generate an initial position output, based on the coarseposition signals, that indicates an initial position of the drive shaftand to generate a subsequent position output, based on the fine positionsignals, that indicates a subsequent position of the drive shaft.

In a particular implementation, a pulse-width modulation control circuitincludes a first transistor and a signal generator. The first transistorincludes a first terminal coupled to a power source and a secondterminal coupled to a first input of a controlled component. The signalgenerator includes a first node coupled to a gate of the firsttransistor. The signal generator is configured to receive a comparisonvalue and a comparison criterion and to compare the comparison value toa counter value based on the comparison criterion. In response to thecomparison value satisfying the comparison criterion with respect to thecounter value, the signal generator is configured to send a controlsignal to the gate of the first transistor to generate a pulse edge of apulse of a pulse-width modulated signal.

In another particular implementation, a system includes a motor and apulse-width modulation control circuit coupled to the motor. Thepulse-width modulation control circuit is configured to output apulse-width modulated signal to the motor. The pulse-width modulationcontrol circuit includes a first transistor and a signal generator. Thefirst transistor includes a first terminal coupled to a power source anda second terminal coupled to a first input of the motor. The signalgenerator includes a first node coupled to a gate of the firsttransistor. The signal generator is configured to receive a comparisonvalue and a comparison criterion and to compare the comparison value toa counter value based on the comparison criterion. In response to thecomparison value satisfying the comparison criterion with respect to thecounter value, the signal generator is configured to send a controlsignal to the gate of the first transistor to generate a pulse edge of apulse of the pulse-width modulated signal.

In yet another particular implementation, a method of pulse-widthmodulation includes receiving a comparison value and a comparisoncriterion and includes comparing the comparison value to a counter valuebased on the comparison criterion. The method further includes, inresponse to the comparison value satisfying the comparison criterionwith respect to the counter value, sending a control signal to a gate ofa first transistor to generate a pulse edge of a pulse of a pulse-widthmodulated signal.

In a particular implementation, feedback control circuitry includes ratelimiter circuitry configured to generate a rate limited position commandbased on a position command for a controlled component and based on aspeed command for the controlled component. The feedback controlcircuitry also includes error adjustment circuitry configured to apply acontrol gain to an error signal to generate an adjusted error signal.The error signal is based on position feedback and the rate limitedposition command, and the position feedback indicates a position of thecontrolled component. The feedback control circuitry further includes anoutput terminal configured to output a current command generated basedon the adjusted error signal.

In another particular implementation, a system includes a motor andfeedback control circuitry coupled to the motor. The feedback controlcircuitry includes rate limiter circuitry configured to generate a ratelimited position command based on a position command for the motor andbased on a speed command for the motor. The feedback control circuitryalso includes error adjustment circuitry configured to apply a controlgain to an error signal to generate an adjusted error signal. The errorsignal is based on position feedback and the rate limited positioncommand, and the position feedback indicates a position of the motor.The feedback control circuitry further includes an output terminalconfigured to output a current command generated based on the adjustederror signal.

In yet another particular implementation, a method for feedback controlincludes receiving a position command for a controlled component and aspeed command for the controlled component and includes generating arate limited position command based on the speed command and theposition command. The method also includes receiving position feedbackindicating a position of the controlled component and applying a controlgain to an error signal to generate an adjusted error signal. The errorsignal is based on the position feedback and the rate limited positioncommand. The method further includes outputting a current command basedon the adjusted error signal.

In a particular implementation, demodulation circuitry includes an inputterminal configured to be coupled to an analog-to-digital converter(ADC) and configured to receive a plurality of ADC outputs. Theplurality of ADC outputs are generated based on resolver outputs. Thedemodulation circuitry also includes a rectifier configured to rectifythe plurality of ADC outputs. Rectifying the plurality of ADC outputspreserves a phase of the plurality of ADC outputs. The demodulationcircuitry includes amplitude determination circuitry configured todetermine, based on the rectified plurality of ADC outputs, demodulatedamplitude values corresponding to the resolver outputs. The demodulationcircuitry further includes angle computation circuitry configured togenerate position outputs based on the demodulated amplitude values.

In another particular implementation, a system includes a resolver, anADC coupled to the resolver, and demodulation circuitry coupled to theADC. The demodulation circuitry is configured to generate demodulatedresolver outputs and includes an input terminal configured to be coupledto the ADC and configured to receive a plurality of ADC outputs. Theplurality of ADC outputs are generated based on resolver outputs. Thedemodulation circuitry also includes a rectifier configured to rectifythe plurality of ADC outputs. Rectifying the plurality of ADC outputspreserves a phase of the plurality of ADC outputs. The demodulationcircuitry includes amplitude determination circuitry configured todetermine, based on the rectified plurality of ADC outputs, demodulatedamplitude values corresponding to the resolver outputs. The demodulationcircuitry further includes angle computation circuitry configured togenerate position outputs based on the demodulated amplitude values.

In yet another particular implementation, a method of demodulatingresolver outputs includes receiving, from an ADC, a plurality of ADCoutputs. The plurality of ADC outputs are generated based on theresolver outputs. The method also includes rectifying the plurality ofADC outputs and rectifying the plurality of ADC outputs preserves aphase of the plurality of ADC outputs. The method includes determining,based on the rectified plurality of ADC outputs, demodulated amplitudevalues corresponding to the resolver outputs. The method furtherincludes generating position outputs based on the demodulated amplitudevalues.

In a particular implementation, dither circuitry includes harmonicsignal generation circuitry configured generate a high order evenharmonic of a base excitation signal. The dither circuitry also includesa combiner configured to generate a dithered excitation signal based onthe high order even harmonic and the base excitation signal. The dithercircuitry further includes an output terminal configured to output thedithered excitation signal to a sensor device.

In another particular implementation, a system includes a resolver, adigital-to-analog converter (DAC) coupled to the resolver, and dithercircuitry coupled to the DAC. The dither circuitry is configured tooutput a dithered excitation signal to the DAC. The dither circuitryincludes harmonic signal generation circuitry configured generate a highorder even harmonic of a base excitation signal. The dither circuitryalso includes a combiner configured to generate the dithered excitationsignal based on the high order even harmonic and the base excitationsignal. The dither circuitry further includes an output terminalconfigured to output the dithered excitation signal to the resolver.

In yet another particular implementation, a method of generating anexcitation signal for a sensor device includes generating a high ordereven harmonic of a base excitation signal. The method further includesgenerating a dithered excitation signal based on combining the highorder even harmonic and the base excitation signal and outputting thedithered excitation signal to the sensor device.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram that illustrates an example of an inertialmeasurement unit;

FIG. 2 is a diagram that illustrates an example of systems of theinertial measurement unit;

FIG. 3 is a diagram that illustrates an example of operation of theinertial measurement unit;

FIG. 4A illustrates a diagram of an example of a transformer;

FIG. 4B illustrates a diagram of an example of a resolver;

FIG. 4C illustrates exemplary graphs of signals of a dual speedresolver;

FIG. 5 is a diagram that illustrates an example of flow processing forprocessing resolver outputs for the dual speed resolver;

FIG. 6 is a logic diagram that illustrates an overview of an example oflogic for processing resolver outputs of the dual speed resolver;

FIG. 7 is a logic diagram that illustrates an overview of logic fordemodulation and angle estimation;

FIG. 8 is a logic diagram that illustrates an example of logic forvoltage conditioning;

FIG. 9 is a logic diagram that illustrates an example of recursivemedian value analysis logic for recursive median value analysis;

FIG. 10 is a logic diagram that illustrates an example of demodulationlogic of FIG. 8;

FIG. 11 is a logic diagram that illustrates an example of logic forrectification with phase preservation;

FIG. 12 illustrates a diagram of exemplary signals generated duringdemodulation;

FIG. 13 illustrates exemplary graphs of signals of the resolver of FIG.4B;

FIG. 14 is a logic diagram that illustrates an example of mask logic ofFIG. 10;

FIG. 15 is a diagram that illustrates an example of masked data andaccumulated data for demodulation;

FIG. 16 is a logic diagram that illustrates an example of output logicof FIG. 10;

FIG. 17 is a diagram that illustrates an example of accumulator inputsand accumulator outputs for demodulation;

FIG. 18 is a logic diagram that illustrates an example of logic forcombining resolver outputs of a dual speed resolver;

FIG. 19 is a logic diagram that illustrates an example of logic for adrift corrector of a dual speed resolver;

FIG. 20 includes a diagram illustrating a mechanical angle estimated bythe resolver system of FIG. 2 and an actual angle of the motor;

FIG. 21 include a diagram that depicts an enlarged view of the diagramof FIG. 20;

FIG. 22A is a diagram that illustrates angles determined based onresolver outputs using an excitation signal without dither;

FIG. 22B is a diagram that illustrates analog-to-digital converter (ADC)outputs generated based on resolver outputs generated by an excitationsignal without dither;

FIG. 23 is a diagram that illustrates an example of a ditheredexcitation signal;

FIG. 24 is a diagram that illustrates ADC outputs generated based on adithered excitation signal;

FIG. 25 is a diagram that illustrates angles determined based on adithered excitation signal;

FIG. 26 is a logic diagram that illustrates an example of logic forexcitation signal generation;

FIG. 27 is a circuit diagram that illustrates an example of a resolverdriver circuit;

FIG. 28 is a circuit diagram that illustrates an example of a motordriver circuit;

FIG. 29 is a diagram that illustrates an example of cascaded feedbacklogic for speed feedback and position feedback;

FIG. 30 is a logic diagram that illustrates an example of logic forcombined speed and position feedback control;

FIG. 31 is a logic diagram that illustrates an example of logic forcombined speed and position feedback control including a direct speedcommand mode;

FIG. 32 is a logic diagram that illustrates an example of logic forcombined speed and position feedback control including an initializationmode;

FIG. 33 is a diagram that illustrates an example of pulse-widthmodulation (PWM) operation with an adjustable comparison criterion;

FIG. 34 is a diagram that illustrates an example of two lane PWMoperation with an adjustable comparison criterion;

FIG. 35 is a logic diagram that illustrates an example of logic for aPWM with an adjustable comparison criterion and dead band control;

FIG. 36 is a flow chart of an example of a method of determiningrotational position using a dual speed resolver;

FIG. 37 is a flow chart of an example of a method of pulse-widthmodulation;

FIG. 38 is a flow chart of an example of a method for feedback control;

FIG. 39 is a flow chart of an example of a method of demodulatingresolver outputs;

FIG. 40 is a flow chart of an example of a method of generating anexcitation signal for a sensor device; and

FIG. 41 is a block diagram that illustrates an example of an aircraftincluding an inertial measurement unit.

DETAILED DESCRIPTION

Implementations disclosed herein are directed to gimbaled inertialmeasurement units. A gimbaled inertial measurement unit includessensors, such as accelerometers and gyroscopes, to determine vehicleinertia data, such as linear acceleration and angular velocity. In agimbaled inertial measurement unit, an inertial measurement unit ismounted on a multi-axis gimbal device. The gimbal device includesmultiple gimbals each with a corresponding motor. The motors are used todrive and position the gimbals such that the sensors are oriented alongthe vehicle's path. A control system of the vehicle tracks the positionof the vehicle based on outputs from the sensors as the gimbals movebased on changes in inertia of the vehicle. The control system thenoutputs commands to the gimbaled inertial measurement unit to adjust(readjust) the sensors such that the sensors are oriented along thevehicle's updated path.

In some implementations, the gimbaled inertial measurement unit uses adual speed resolver for determining a position of the motor (e.g., adrive shaft of the motor) and thereby a position the sensors attached tothe corresponding gimbal. The dual speed resolver determines theposition of the drive shaft of the motor using two resolvers each havinga different “speed”. A first resolver (e.g., a coarse resolver) may havea first speed that corresponds to a speed and a position (e.g., anabsolute position) of the drive shaft. A resolver speed corresponds to anumber of electrical cycles (e.g., a sine wave or a cosine wave)generated by a single mechanical revolution of the resolver. In aparticular implementation, the first resolver is driven or rotated atthe same rotational speed as the drive shaft, and thus an electricalcycle of the first resolver corresponds to a mechanical revolution ofthe drive shaft. Accordingly, the absolute position of the drive shaftcan be determined from the first resolver.

A second resolver (e.g., a fine resolver) has a second speed that isgreater than the first speed and that corresponds to a position of thedrive shaft. For example, the second resolver may include multiple poles(e.g., pairs of coils) which generate multiple electrical cycles (e.g.,sine waves) for a single mechanical revolution of the resolver (and thedrive shaft). Alternatively, the second resolver may complete multiplemechanical revolutions for each revolution of the drive shaft. Ascompared to the first resolver (e.g., the coarse resolver), the secondresolver (the fine resolver) has increased precision at the expense ofnot being able to determine a starting position (e.g., an absolutestarting position). The second resolver can determine a more preciselocation of the drive shaft, but is unable to determine in whichquadrant of a 360 degree rotation the drive shaft is in.

Dual speed resolvers (or dual resolvers) use outputs of both resolversto determine a position of the drive shaft. For example, in conventionaldual speed resolvers, outputs of both resolvers are input into a KalmanFilter to increase precision over a single resolver. However, the outputof the coarse resolver has less accuracy and precision than the fineresolver and utilizing both the coarse and fine outputs reduces theaccuracy and precision of the dual resolver to less than the accuracy ofthe fine resolver. By using the coarse resolver outputs to determine astarting position (e.g., during an initialization process or timeperiod) and the fine resolver outputs to determine subsequent positions(e.g., positions after the initialization process or time period), theaccuracy and precision of the dual speed resolver is increased overconventional dual speed resolvers. Such a dual speed resolver can beused to determine the absolute starting position and have the accuracyand precision of the fine resolver. Additionally, the fine resolveroutputs can further be used to correct for a starting offset (error) ofthe coarse resolver. Furthermore, other methods described herein can becombined to further increase the accuracy and precision of the dualspeed resolver.

As explained above, a resolver receives excitation signals and inresponse generates an output signal. By adding dither (e.g., zero meandither) to the excitation signal, the precision of the angle determinedfrom the resolver output is increased without increasing a speed of theresolver (e.g., a number of poles of the resolver) or a bandwidth (e.g.,a sampling frequency of outputs of the resolver or a number of bits ofthe resolver outputs) of the processing circuitry. Zero mean ditherincludes or corresponds to noise that does not alter a median amplitudevalue of the base excitation signal. Additionally, by time coordinatingthe dithered excitation signal with current drive switching signals ofthe gimbal motors, the dithered excitation signal can produce outputsthat have less noise and interference. Accordingly, the precision of theangle determined from the resolver output is increased withoutincreasing a speed of the resolver or a bandwidth of the processingcircuitry.

As explained above, resolver outputs are converted to digital samples byan analog-to-digital converter (ADC) and the digital samples aredemodulated during processing to determine the angle of the resolver andthe drive shaft. During demodulation, the digital samples are rectifiedto produce a rectified signal. In some implementations, the digitalsamples are rectified such that a phase of the excitation signal ispreserved in the rectified signal. For example, conventionaldemodulators multiply the digital samples by the excitation signal topreserve the phase of when rectifying the digital samples. However,multiplying the digital samples by the excitation signal generatesnoise. To illustrate, multiplying sine values of angles together reducesaccuracy of the data between peaks of the sine waves, i.e., it squaresany noise or errors.

Rectifying the digital samples with a square wave improves a signal tonoise ratio of off the data between peaks of the sine waves i.e., offpeak voltages. Additionally, by flipping a sign of the square wave inaccordance with the phase of the excitation signal, the phase of thedigital samples and excitation signal can be preserved without impartingadditional noise or reducing the signal to noise ratio. Accordingly, theprecision of the angle determined from the resolver output is increasedwithout increasing a speed of the resolver or a bandwidth of theprocessing circuitry.

Additionally, recursive median value analysis and masking portions ofthe data further increases precision during demodulation. Recursivemedian value analysis may be applied to the input digital samples and tothe output amplitudes of the demodulator to further increase precision.For example, the demodulator may use a median value (a midvalue) of thelast n input samples as the input value, where n is any integer greaterthan 1. As another example, the demodulator may output a midvalue of thelast m output samples, where m is any integer greater than 1.Additionally or alternatively, the demodulator may output a midvalue of3 different signals as the output value. The output value is used todetermine the angle of the resolver and the drive shaft.

In some implementations, the demodulator masks portions of the data toeliminate noise and interference, which further increases precision ofthe gimbaled inertial measurement. The masked portion includes noisydata (data occurring during current drive switching and includingcurrent drive interference), data corresponding to a transition betweenpeak amplitudes, or both. Thus, the demodulation improves results byusing data near peak amplitudes of the excitation signal. Additionallyor alternatively, demodulation outputs are based on a synced (syncedwith the excitation signal) accumulator output to further increaseprecision and reduce errors. For example, by synching the accumulatorwith the excitation signal that is time coordinated with current drivesignals, the outputs of the accumulator can mask at least a portion ofthe current drive interference, average out the effects of the currentdrive interference, or both, leading to increased precision andaccuracy.

The gimbaled inertial measurement unit also includes a feedback controlsystem to control the gimbal motors. Gimbal motors are usually commandedor controlled by a direct rate command or a position command and a ratecommand. These commands may be received from user input or a vehicle'scontroller (e.g., a flight computer). Feedback control systems generallyuse a cascaded (e.g., multi-loop) tracking control law to process theposition command and the rate command and to provide feedback. By usinga combined rate and position feedback system (e.g., a single loopfeedback system), lightly damped gimbal motors can be used with tightrate gains to achieve greater precision.

A pulse-width modulator (PWM) is used to drive the gimbal motors. ThePWM controls activation of the gimbal motors based on the feedbackcontrol system. For example, when the gimbal motors correspond to3-phase motors, the PWM controls the power delivery to the gimbal motorsbased on current commands generated by the feedback control system. Toillustrate, the current commands are converted in a duty cycle value orsignal. For example, the current commands are indicative of an amount ofcurrent to be provided to the motor. The duty cycle value (e.g., 50percent) is determined based on the amount of current and a voltage ofthe power supply or motor. The duty cycle signal (e.g., a set pointsignal) indicates the duty cycle value. To illustrate, for an 8 bit dutycycle signal a value of 31 or 32 may indicate 50 percent duty cycledepending on which comparison condition is used. The duty cycle signalis sent to the PWM which generates pulses. A width of the pulsescontrols power delivery to the gimbal motors.

The PWM generates the pulses based on comparing a counter value to acomparison value (indicative of a duty cycle value, such as 50 percent,51 percent, etc.). For example, the PWM generates pulses based ondetermining whether the counter value is greater than or less than thecomparison value. To illustrate, the PWM generates a first pulse edge(e.g., activates a gate of a transistor) of a pulse when the countervalue is greater than the comparison value and generates a second pulseedge (e.g., deactivates the gate of the transistor) of the pulse whenthe counter value is no longer greater than (does not exceed) thecomparison value. In conventional PWMs, increasing a precision orreducing granularity of control adjustment requires increasing anoperating frequency of PWM components.

By utilizing an adjustable comparison criterion, PWM precision ofcontrol can be increased and control granularity can be reduced withoutincreasing the operating frequency of the PWM components. The adjustablecomparison criterion may be indicated by a set point signal. Theadjustable comparison criterion includes other comparison conditions orrules, such as a greater than or equal to condition and a less than orequal to condition. PWM's utilize up-down counters which generate acounter signal that is a triangle wave. Thus, when adjusting thecomparison value by one counter value (clock pulse), the PWM generate afirst pulse one clock pulse earlier and a second pulse one clock pulselater, leading to an increase in pulse-width of two clock pulses.However, when adjusting the comparison criterion, the PWM can generate afirst pulse one clock pulse earlier and a second pulse at the same time,leading to an increase in pulse-width of one clock pulse. Accordingly,by using an adjustable comparison criterion, a PWM has increasedprecision and reduced granularity of control adjustment of the motorwithout increasing an operating frequency of PWM components.

By utilizing one or more of the above improvements, a gimbaled inertialmeasurement unit can offer greater precision in a smaller footprint overconventional gimbaled inertial measurement units. Additionally, thegimbaled inertial measurement unit can achieve the increased precisionwithout increasing hardware capability. Accordingly, the gimbaledinertial measurement unit enables a vehicle to inertial navigate saferdue to the increased precision. Additionally, vehicles using thegimbaled inertial measurement unit may be smaller and lighter, leadingto lower costs.

FIG. 1 is a diagram 100 that illustrates an example of an inertialmeasurement unit 102, such as a gimbaled inertial measurement unit. Insome implementations, the inertial measurement unit 102 is included in avehicle (e.g., a ship, a submarine, an aircraft, a rocket, a satellite,a spacecraft, etc.) and is coupled to control system thereof, as shownin FIGS. 2 and 41.

The inertial measurement unit 102 includes an inverter 112 and a gimbaldevice 114. The inverter 112 includes inverter electronics and firmware.The inverter 112 is configured to receive direct current (DC) power,convert the DC power to alternating current (AC) power, and provide theAC power to the gimbal device 114. For example, the inverter 112 isconfigured to provide power to control operation of the motors 124 ofthe gimbal device 114. The inverter electronics may include orcorrespond to a processor, a field programmable gate array (FPGA), anapplication specific integrated circuit (ASIC), or a combinationthereof. The firmware is configured to control operation of the inverterelectronics. In some implementations, the inverter electronics andfirmware include or correspond to a PWM, such as the PWM 242 of FIG. 2,configured to control power delivery (e.g., current drive switching) tothe motors 124.

The gimbal device 114 includes or corresponds to a multi-axis gimbal ora set of gimbals 122. The gimbal device 114 is configured to point thesensors 128 relative to a path of the vehicle. In some implementations,the gimbal device 114 includes a three axis gimbal. In a particularimplementation, the three axis gimbal includes a ball (e.g., a firstaxis gimbal 122), an inner shell (e.g., a second axis gimbal 122), andan outer shell (e.g., a third axis gimbal 122). In otherimplementations, the gimbal device 114 includes a two gimbal system or afour gimbal system.

The gimbal device 114 includes the motors 124. Each motor 124 isconfigured to drive or control a position of a corresponding gimbal 122of the gimbal device 114 to orient the sensors 128 attached to thecorresponding gimbal 122 in line with the path of the vehicle. Themotors 124 may include or correspond to electric motors, such as abrushed electric motor or brushless electric motor.

The gimbal device 114 includes multiple resolvers 126. Each resolver 126is configured to determine a position of a corresponding gimbal 122 ofthe gimbal device 114 such that the sensors 128 attached to thecorresponding gimbal 122 can be oriented in line with the path of thevehicle. For example, each resolver 126 is coupled to a drive shaft of acorresponding motor 124 and outputs of each resolver 126 are used todetermine a position of the drive shaft, and thus the position of thecorresponding gimbal 122 and set of sensors 128.

The sensors 128 include accelerometers 132 and gyroscopes 134. Theaccelerometers 132 are configured to determine linear acceleration andthe gyroscopes 134 are configured to determine angular velocity. Forexample, the accelerometers 132 and gyroscopes 134 generate sensor dataindicative of linear acceleration and angular velocity or changes inlinear acceleration and angular velocity. In some implementations, eachgimbal 122 includes a set of sensors 128. Each set of sensors 128includes one or more accelerometers 132 and one or more gyroscopes 134.

In some implementations, the sensors 128 further include magnetometers.In a particular implementation, each gimbal 122 of the gimbal device 114further includes a magnetometer. The magnetometers are configured todetect a direction, a strength, or relative change of a magnetic field.Outputs of the magnetometers can be used to determine heading and/orposition of the vehicle.

During operation of the vehicle, the vehicle may change its direction(e.g., a path, heading, or course). For example, the vehicle may changeits direction from a first direction (e.g., an original direction) to asecond direction (e.g., an updated direction). In response to thevehicle changing directions to the second direction, the inertialmeasurement unit 102 positions (e.g., repositions) the gimbals 122 ofthe gimbal device 114 to orient (e.g., point) the sensors 128 attachedto the gimbals 122 along the second direction (e.g., the vehicle'scurrent path or heading). The inertial measurement unit 102 positionsthe gimbals 122 of the gimbal device 114 based on the sensor datagenerated by the sensors 128 when the vehicle changed from the firstdirection to the second direction.

The vehicle may change its direction again. For example, the vehicle maychange its direction from the second direction to a third direction. Inresponse to the vehicle changing direction for the second time, theinertial measurement unit 102 positions (e.g., repositions) the gimbals122 of the gimbal device 114 to orient (e.g., point) the sensors 128attached to gimbals 122 along the third direction (e.g., the vehicle'scurrent path). The inertial measurement unit 102 positions the gimbalsof the gimbal device 114 based on the sensor data generated by thesensors 128 when the vehicle changed from the second direction to thethird direction. Although, the above examples utilize a change indirection, the gimbal device 114 can sense any change in inertia (suchas a change in speed along the same direction or heading) of the vehicleand the inertial measurement unit 102 can position the gimbal device 114responsive to the change in inertia of the vehicle. The inertialmeasurement unit 102 and components thereof are described further withrespect to subsequent figures.

FIG. 2 illustrates a diagram 200 of an example of systems of theinertial measurement unit 102 of FIG. 1. In the diagram 200, the gimbaldevice 114 is not shown for clarity. In the particular exampleillustrated in FIG. 2, the inertial measurement unit 102 includes anexcitation signal generation system 202, a resolver system 204, and acontrol system 206. Each of these systems, or subsystems thereof,improve the function of the inertial measurement unit 102 individuallyand in combination with the other systems, as will be described in moredetail in subsequent figures. Additionally, the inertial measurementunit 102 includes the inverter 112 and the motors 124 described withreference to FIG. 1.

The inertial measurement unit 102 and components thereof may be coupledto other equipment of the vehicle. As illustrated in FIG. 2, theinertial measurement unit 102 and components thereof are coupled to apower supply 252 and a flight computer 254. In some implementations, thepower supply 252 corresponds to a DC power supply of the vehicle, suchas a battery or a generator. In other implementations, the power supply252 may be included in the inertial measurement unit 102, e.g., as aninternal battery. The flight computer 254 may include or correspond to aflight control computer (FCC) or a guidance system configured to controlthe vehicle, such as cause a change in the course or direction of thevehicle responsive to user input or autonomously.

The excitation signal generation system 202 is configured to generateexcitation signals and to output the excitation signals to the resolversystem 204. The excitation signals are configured to cause the resolvers126 to generate output signals indicative of a position of a drive shaftof the corresponding motor 124, as described further with reference toFIG. 4.

The excitation signal generation system 202 includes a dither generator212 and coordination system 214. The dither generator 212 is configuredto generate and add dither to the excitation signal (base excitationsignal) to generate a dithered excitation signal. The ditheredexcitation signal enables more precise demodulation and more precisegimbal/motor control, which leads to better sensor outputs and increasedprecision of the inertial measurement unit 102, as described furtherwith reference to FIGS. 22-29.

The coordination system 214 is configured to coordinate the excitationsignal (or dithered excitation signal) with current drive switching ofthe motors 124 to generate a coordinated excitation signal (orcoordinated dithered excitation signal). To illustrate, turning on andoff transistors which provide power to the motors 124 may generatespikes, positive and negative spikes respectively. Waves of theexcitation signal may be coordinated with (e.g., offset from) currentdrive switching such that a same number of transitions from off to onand from on to off occur during each wave. Additionally, the currentdrive switching may be offset from peak amplitudes of the waves of theexcitation signal. The coordinated excitation signal reduces noise andcontamination of resolver outputs, which leads to better sensor outputsand increased precision of the inertial measurement unit 102.

The resolver system 204 is configured to generate resolver outputsindicative of a position of the drive shafts of the motors 124responsive to the excitation signals. The resolver outputs are analogsignals which are converted to digital samples by an ADC and processedto determine a position (an angle) of the drive shaft. The resolversystem 204 includes the resolvers, 126, a demodulation system 222, and adual resolver combination system 224.

The demodulation system 222 is configured to demodulate the digitalsamples output by the ADC, as described further with reference to FIGS.7-17. In some implementations, the demodulation system 222 performsrecursive median value analysis to reduce or eliminate spikes in thedigital samples input to the demodulation system 222, such as spikescaused by current drive switching and other interference. Additionallyor alternatively, the demodulation system 222 includes an accumulator toreduce or eliminate spikes in the demodulation output of thedemodulation system 222. The accumulator may also preform recursivemedian value analysis and masking (e.g., filtering) noisy data to reduceor eliminate spikes in the demodulation output.

The dual resolver combination system 224 is configured to generate angleestimations based on the demodulation outputs and to combine angleestimations from each resolver to determine the positions of the driveshafts, as described further with reference to FIGS. 18 and 19. The dualresolver combination system 224 uses the coarse resolver to determine astarting position of the drive shaft during an initialization process(e.g., an initialization mode). The starting position corresponds to aninitial position of the drive shaft in absolute terms (0 to 360degrees). The dual resolver combination system 224 uses the moreaccurate and precise fine resolver outputs for determining subsequentpositions of the drive shafts after the initialization process. In someimplementations, the dual resolver combination system 224 includes adrift corrector to correct for drift. The drift corrector may use thefine resolver outputs to correct for an initial error (offset) of thestarting position determined by the coarse resolver and to correct forintegration errors of the fine resolver outputs.

The control system 206 includes a feedback control system 232. Thefeedback control system 232 is configured to receive flight controlinputs from the flight computer 254, position feedback from the resolversystem 204, and speed feedback (e.g., revolutions per minute (RPM)feedback) from the resolver system 204. The position feedback isindicative of a position (angle) of the motors 124 and the speedfeedback is indicative of a rate, such as a rate of the motor in RPM.

The flight control inputs include a speed command (e.g., RPM command), aposition command, or both. The feedback control system 232 generates acurrent command based on the flight control inputs, the positionfeedback, and the speed feedback, as described further with reference toFIGS. 30 and 31. The control system 206 converts the current commandinto a duty cycle setting or value used to control the motors 124. Insome implementations, the duty cycle setting is indicated by a set pointsignal that indicates a comparison value and a comparison criterion.

The inverter 112 includes a PWM 242 configured to apply adjustablecomparison criteria. The PWM is configured to receive the comparisonvalue and one or more comparison criteria and to generate pulses of apulse-width modulated signal based on the comparison value and one ormore comparison criteria, as described with reference to FIGS. 33-35.The pulse-width modulated signal has increased precision or reducedgranularity of control and is used to more precisely provide power fromthe power supply 252 to the motors 124. Operation of the inertialmeasurement unit 102 of FIG. 2 is described with reference to FIG. 3.

FIG. 3 is a diagram 300 that illustrates an example of operation of theinertial measurement unit 102. Diagram 300 corresponds to operation of asingle motor 124 of the inertial measurement unit 102 and whichpositions the sensors 128 of a particular gimbal 122 (corresponding tothe motor 124) of the gimbal device 114. In FIG. 3, cross hatchingdenotes operations or steps that may be time coordinated with eachother. The time coordination can be achieved by using the same clock orcounter, by synchronizing two or more clocks or counters, by offsettingtwo or more clocks or counters, or a combination thereof, as describedfurther with reference to FIGS. 27 and 28.

During operation of the inertial measurement unit 102, the excitationsignal generation system 202 generates a dithered excitation signal 352,coordinates the dithered excitation signal with other components of theinertial measurement unit 102 (such as one or more cross hatchedcomponents), and provides the dithered excitation signal 352 to adigital-to-analog converter (DAC) 310. The DAC 310 converts the ditheredexcitation signal 352 into an analog signal and provides the analogdithered excitation signal 352 to each resolver 342, 344 of the dualspeed resolver 312. For example, the dual speed resolver 312 includes a1 speed resolver and 16 speed resolver. Each resolver 342, 344 generatesan output, such as a differential voltage output 354. For example, asillustrated in FIG. 3, the coarse resolver 342 outputs differentialvoltage output 354A (i.e., coarse position signals indicative of acourse position) and the fine resolver outputs differential voltageoutput 354B (i.e., fine position signals indicative of a fine position).

The differential voltage outputs 354 are measured by correspondingdifferential voltage sensors 314, 316 to generate differential voltagesignals 356. The differential voltage signals 356 generated by thedifferential voltage sensors 314, 316 are sampled by corresponding ADCs318, 320. The ADCs 318, 320 output digital samples of voltage values(referred to as ADC outputs 358) to voltage conditioning circuitry 322,324 which correct for a voltage bias of the inertial measurement unit102. The conditioned voltage values 360 are demodulated by thedemodulation system 222 to generate demodulated outputs 362. Angleestimation circuitry 326 generates angle estimates 364 for bothresolvers from the demodulated outputs 362. The angle estimates 364 areused to generate an estimated position 366 of the motor, such as aninitial position and subsequent positions of the motor, based on thedemodulated outputs 362. One or more estimated positions 366 are used todetermine an estimated RPM 368 of the motor. Additionally, the estimatedpositions 366 may be adjusted (tared) to account for the motor 124, suchas to account for commutation and servo offset. Adjusting (taring) theestimated positions 366 generates a magnetic rotor position and amechanical rotor position, such as tared position outputs. The estimatedRPM 368 may be determined based further on the magnetic rotor positionand the mechanical rotor position. The estimated position 366 and theestimated RPM 368 of the motor 124 are provided to the feedback controlsystem 232 to be used as feedback, i.e., position feedback and RPMfeedback, for controlling the motor 124.

The feedback control system 232 receives commands 370 from the flightcomputer 254, such as a position command and an RPM command. Thefeedback control system 232 generates a rate limited position commandbased on the position command and the RPM command. The feedback controlsystem 232 generates a current command 372 based on the rate limitedposition command, the position feedback, and the RPM feedback. Thecurrent command 372 may correspond to a torque command or indicate anamount of torque of the motor 124. The current command 372 is providedto a current tracker 330 to generate a duty cycle value 374. The dutycycle value 374 may be in terms of a two phase reference frame (e.g., atwo phase reference frame of direct and quadrature, with quadraturecorresponding to torque). An inverse Park/Clark transformation 332 canbe applied to the duty cycle value 374 to convert the duty cycle value374 to one or more duty cycle settings 376 for a three phase motor, suchas a duty cycle in terms of A, B, and C lanes (phases) of the threephase motor. The one or more duty cycle settings 376 (e.g., set pointsignals) are sent to the PWM 242.

The PWM 242 receives the one or more duty cycle settings 376, each dutycycle setting 376 indicating a comparison value and two comparisoncriteria. The PWM 242 may receive one duty cycle setting 376 for eachlane (A, B, and C) or may receive one duty cycle setting 376 for aparticular lane and generate duty cycle settings 376 for each other lanebased on the received duty cycle setting 376. The PWM 242 controls gatedrivers 334 of transistors 336 (metal-oxide-semiconductor field-effecttransistors (MOSFETs)) of the inverter 112 based on the comparing thecomparison value to a counter value based on the two comparison criteriafor each lane.

The PWM 242 is configured to generate (or cause the gate drivers 344 togenerate) pulse-width modulated signals 378. The PWM 242 generatespulses (a pulse for each comparison criteria) of the pulse-widthmodulated signals 378 for each lane, and the pulses activate anddeactivate the transistors 336 (i.e., current drive switching). Byactivating the gate drivers 334, the PWM 242 controls power deliveryfrom the power supply 252 to the motors 124. The PWM 242 and theinverter 112 convert the DC power from the power supply 252 into threephase AC power signals 380 with increased precision for controlling themotors 124.

FIGS. 4A-4C illustrate example operation of a resolver. FIG. 4Aillustrates a diagram of an example of a transformer 402, two coils 412,414 of wire (referred to as “windings”) wrapped around a magnetic core.A change in current applied to an input coil 412 creates a varyingmagnetic flux and varying magnetic field at an output coil 414. Thevarying magnetic field at the output coil 414 induces voltage in theoutput coil 414. A resolver operates using a rotatable transformer andoutputs induced voltage.

FIG. 4B illustrates a diagram of an example of a resolver 404. Theresolver 404 is an analog sensor configured to determine rotationalposition of a rotating component. The resolver 404 is an active sensor,i.e., it receives an excitation signal, such as the dithered excitationsignal 352 of FIG. 3, which causes (induces) an output signal.

The resolver 404 includes three coils 422-426 forming a rotatabletransformer. The resolver 404 includes a rotatable primary coil 422(first coil) and two secondary coils 424, 426 (sine and cosine coils).The secondary coils 424, 426 may include or correspond to pairs ofpoles, e.g., 2n poles. Each pole is angularly offset from the otherpole, such as by 90 degrees. Each pair of poles has a pole configured todeliver a sine output and another pole configured to deliver a cosineoutput. As the primary coil 422 rotates, the primary coil 422 receivesthe excitation signal 352 and induces voltage in the secondary coils424, 426. Each of the secondary coils 424, 426 may output a differentialoutput, such as the differential voltage outputs 354 of FIG. 3. Theratio between the voltages of the secondary coils 424, 426 represents anangle 430 of the resolver 404 (which indicates an angle of the motor124). The resolver 404 in FIG. 4B is a two pole resolver and is a singlespeed resolver, e.g., the coarse resolver 342 of FIG. 3.

In multispeed resolvers, the multispeed resolver (e.g., the fineresolver 344) has a higher number of electrical cycles per rotation ofthe multispeed resolver (and the component being tracked). For example,the multispeed resolver can be a multipole resolver and extra pairs ofpoles (coils) can be added to the resolver 404 to produce moreelectrical cycles per one mechanical rotation of the primary coil 422(and the component being tracked). As another example, the primary coil422 of the multispeed resolver can be geared relative to the componentbeing tracked such that one mechanical rotation for of the componentbeing tracked causes more than one rotation of the primary coil 422.

FIG. 4C illustrates exemplary graphs 462-468 of signals of the dualspeed resolver 312 of FIG. 3. Each of the graphs 462-468 of FIG. 4Ccorresponds to the same interval of time, which corresponds to a partialrotation of the resolvers 342, 344 of the dual speed resolver 312. Afirst graph 462 depicts voltages of an excitation signal 452 (an ACsignal) over the interval of time supplied to both resolver 342, 344. Asillustrated in FIG. 4C, the excitation signal 452 has a frequency of2442 Hertz. In other implementations, the excitation signal is adithered excitation signal, such as the dithered excitation signal 352of FIG. 3 and illustrated in FIG. 23. The excitation signal 452 (baseexcitation signal) is depicted in FIG. 4 for illustrative purposes.

A second graph 464 depicts voltage outputs of the sine secondary coil424 of the coarse resolver 342 (e.g., the resolver 404) over theinterval of time. A third graph 466 and a fourth graph 468 depictvoltage outputs of the fine resolver 344 over the interval of time. Asillustrated in FIG. 4C, the fine resolver 344 is a 16 speed resolver andthe third graph 466 depicts voltage outputs of the sine secondary coil424 and the fourth graph 468 depicts voltage outputs of the cosinesecondary coil 426 of the fine resolver 344. As compared to the voltageoutputs of the sine secondary coil 424 of the coarse resolver 342 of thesecond graph 464, the voltage outputs of the sine secondary coil 424 ofthe fine resolver 344 of the third graph 466 have higher voltages, ahigher cycle frequency, and complete multiple electrical cycles over theinterval of time. As compared to the voltage outputs of the sinesecondary coil 424 of the fine resolver 344 of the third graph 466, thevoltage outputs of the fine resolver 344 of the cosine secondary coil426 of the fourth graph 468 have the same cycle frequency but are offset(e.g., out of phase) relative to the third graph 466. The voltageoutputs of the cosine secondary coil 426 of the fine resolver 344 of thefourth graph 468 are also offset from the excitation signal 452 of thefirst graph 462. Processing of resolver outputs, e.g., demodulation andangle combination, are described with reference to FIGS. 5-21.Generation of excitation signals for the resolver 404 are described withreference to FIGS. 22-28.

FIG. 5 is a diagram 500 that illustrates an example of flow processingfor processing resolver outputs for the dual speed resolver 312 of FIG.3. As illustrated in FIG. 5, the resolver system 204 includes twoprocessing chains 502, 504, one for each resolver of the dual speedresolver 312. The processing chains 502, 504 include the ADCs 318, 320,demodulation circuitry 514, 524, and angle calculation circuitry 516,526. Outputs from each processing chain 502, 504 are input into outputcircuitry 532.

As illustrated in FIG. 5, the output circuitry 532 includes anglecombination circuitry 542 and drift correction circuitry 544. A positionof the drive shaft of the motor can be determined based on outputs fromthe angle combination circuitry 542 and the drift correction circuitry544.

During operation, the coarse resolver 342 (e.g., a first resolver or onespeed resolver) of FIG. 3 outputs coarse position signals to an input ofthe first ADC 318, and the fine resolver 344 (e.g., a second resolver ora multi-speed resolver) of FIG. 3 outputs fine position signals to aninput of the second ADC 320. In the example illustrated in FIG. 5, theADCs 318, 320 receive the differential voltage signals 356 of FIG. 3. Ina particular implementation, the fine position signals are 16 speed. Thecoarse and fine position signals include analog sine and cosine waves.In a particular implementation, the coarse and fine position signalseach include a differential sine signal and a differential cosinesignal.

The ADCs 318, 320 convert the analog outputs of the resolvers intodigital samples. The digital samples output by the ADCs 318, 320 arereceived by the corresponding demodulation circuitry 514, 524. Thedemodulation circuitry 514, 524 demodulates the digital samples toproduce amplitudes of the sine and cosine waves. In a particularimplementation, the amplitudes include sign information, i.e., are“signed” and indicate whether the sample is positive or negative.Details of the demodulation are described further with reference toFIGS. 6-17.

The amplitude outputs of the demodulation circuitry 514, 524 arereceived by the corresponding angle calculation circuitry 516, 526. Theangle calculation circuitry 516, 526 calculates estimated angles of thedrive shaft using the sine and cosine amplitudes. In a particularimplementation where the amplitudes include sign information, the anglecalculation circuitry 516, 526 calculates the estimated angles of theresolvers 342, 344 (which are indicated of angles of the drive shaft)using arctan 2 (commonly abbreviated as atan 2). The atan 2 function isa four quadrant inverse arc tangent function capable of determiningwhich quadrant the drive shaft is in based on the signed amplitudes.

The estimated angles of the drive shaft from each of the anglecalculation circuitry 516, 526 are provided to output circuitry 532. Inthe example illustrated in FIG. 5, the angle combination circuitry 542receives both estimated angle outputs, and the drift correctioncircuitry 544 receives the multispeed estimated angle output.

The angle combination circuitry 542 combines the estimate angles (e.g.,the angle estimates 364 of FIG. 3) from each angle calculation circuitry516, 526 to determine a starting position (an initial estimated position366 of FIG. 3) of the drive shaft and subsequent positions of the driveshaft. In a particular implementation, the angle combination circuitry542 uses estimated angle outputs from the first angle calculationcircuitry 526 to determine the starting position and uses estimatedangle outputs from the second angle calculation circuitry 526 todetermine the subsequent positions of the drive shaft. Alternatively,the angle combination circuitry 542 uses both estimated angle outputs todetermine the starting position and uses estimated angle outputs fromthe second angle calculation circuitry 526 to determine the subsequentpositions of the drive shaft.

The drift correction circuitry 544 corrects for errors that are producedwhen combining the two estimated angles. For example, when determiningthe starting and subsequent positions of the drive shaft includesdifferentiation of the multispeed estimated angles, noise and integererror may be introduce into outputs of the angle combination circuitry542. To correct for the noise and integer error, the angle combinationcircuitry 542 receives, an estimated position from the angle combinationcircuitry 542. The drift correction circuitry 544 generates a driftcorrection output based on the multispeed estimated angles from thesecond angle calculator. The drift correction circuitry 544 provides thedrift correction output to the angle combination circuitry 542. Theangle combination circuitry 542 adjusts subsequent outputs (subsequentestimated positions 366 of FIG. 3) based on the drift correction output.Details of the angle combination circuitry 542 are described furtherwith reference to FIGS. 6, 7, and 18, and details of the driftcorrection circuitry 544 are described further with reference to FIGS.6, 7, and 19.

FIG. 6 is a logic diagram 600 that illustrates an overview of an exampleof logic 602 for processing resolver outputs of the dual speed resolver312 of FIG. 3. The logic diagram 600 depicts an overview of conversionof resolver outputs to ADC outputs, demodulation of the ADC outputs,angle estimation based on the demodulated outputs, and a combinedposition output based on the angle estimations. Each circled portion ofthe logic diagram 600 is described in further detail with respect toFIGS. 7-19. Logic or portions thereof of FIG. 6 and subsequent FIGS. maybe performed by one or more application specific circuits (circuitry),an FPGA, firmware (such as firmware of an FPGA), software executed by aprocessor, or a combination thereof. Additionally, the logic (orportions thereof) disclosed herein may be replaced by equivalent logic.For example, any logic gate can be represented by one or more NOR logicgates.

FIG. 7 is a logic diagram 700 that illustrates an overview of an exampleof logic 702 for demodulation and angle estimation. The logic 702depicts demodulation and angle estimation for a single resolver of thedual speed resolver 312. As illustrated in FIG. 7, the logic 702corresponds to logic for demodulation and angle estimation for the fineresolver 344 (e.g., 16 speed) of FIG. 3.

The logic 702 includes demodulation logic 712 and angle estimation logic714. The demodulation logic 712 is configured to the receive theresolver outputs (e.g., the differential voltage outputs 354) via theADC 320 as the ADC outputs 358 and to output the demodulated outputs 362to the angle estimation logic 714, as described further with referenceto FIGS. 8-17. The angle estimation logic 714 is configured to receivethe demodulated outputs 362 (demodulated amplitude values) and calculatethe angle estimates 364 for the resolvers, as described further withreference to FIGS. 18 and 19.

In some implementations, the angle estimation logic 714 is configured tocalculate the angle estimates 364 based on a function of atan 2 (e.g.,four quadrant inverse arc tangent). For example, the angle estimationlogic 714 is configured to generate angle estimates 364 based on aproduct of the demodulated outputs 362 (demodulated amplitude values)and based on sine and cosine feedback angle values 742, 744.

By calculating the angle estimates 364 based on a function of atan 2,the angle estimates 364 indicate which quadrant the motor is in. Toillustrate, an angle output by atan 2 is from 0-360 degrees (as opposedto 0 to 90 degrees). The function atan 2 uses signed input angles todetermine the specific quadrant (i.e., 0-90, 90-180, 180-270, or 270-360degrees).

In some implementations, the demodulation logic 712 is furtherconfigured to adjust (tare) for voltage bias and other biases of thehardware, as described further with reference to FIG. 8. In someimplementations, the angle estimation logic 714 is further configured tooutput estimated angle outputs for the sine and cosine values inaddition to the angle estimates 364.

During operation, the demodulation logic 712 receives digital samplesfrom an ADC. For example, the demodulation logic 712 receives the ADCoutputs 358 from the one of the ADCs 318, 320. The digital samplesrepresent voltages of the sine and cosine coils 424, 426 of theresolvers 342, 344 of the dual speed resolver 312. As illustrated inFIG. 7, the digital samples correspond to sine and cosine inputs fromthe fine resolver 344. The digital samples are demodulated to producedemodulated sine and cosine outputs 362. Multiple demodulated outputs362 may be referred to as demodulated feedback or a demodulated feedbacksignal. In a particular implementation, the demodulation logic 712generates a median value of the past three demodulated outputs 362 andoutputs the median value as next the demodulated output 362 of thedemodulated feedback signal. Detailed operation of the demodulationlogic 712 is described further with reference to FIGS. 8-16.

The demodulated outputs 362 are multiplied by the sine and cosinefeedback angle values 742, 744 to generate products 752, 754. Toillustrate, the sine demodulated outputs 362 are multiplied by thecosine feedback angle values 744 to generate first products 752. Thecosine demodulated outputs 362 are multiplied by the sine feedback anglevalues 746 to generate second products 754. The sine and cosine feedbackangle values 742, 744 may be generated by the angle estimation logic 714as sine and cosine estimated angle values 762, 764 and provided to thelogic 702. For example, the sine and cosine estimated angle values 762,764 are generated based on the demodulated outputs 362. To illustrate,the sine estimated angle values 762 are calculated based on the firstproduct 752 of the sine feedback angle values 742 and values(amplitudes) of the sine demodulated outputs 362.

The angle estimation logic 714 receives the products 752, 754 of thedemodulated outputs 362 and the sine and cosine feedback angle values742, 744 and generates the angle estimates 364 based on the demodulatedoutputs 362. For example, the angle estimation logic 714 generates theangle estimates 364 based on the function atan 2 (i.e., four quadrantinverse arc tangent). To illustrate, a particular angle estimate 364 iscalculated by applying the function atan 2 to a value (angle) of anintegral of the product 752 minus the product 754, i.e., integral (sineamplitude*cos(θ)−cosine_amplitude*sin(θ)−0). In such implementations,the demodulated outputs 362 are signed, i.e., include sign information.As explained with reference to FIGS. 7, 18, and 19, the angle estimates364 indicated by the fine resolver 344 may be used in combination withthe angle indicated by the coarse resolver 342 to generate an initialposition and subsequent positions of the drive shaft of the motor 142.

In some implementations, the demodulation logic 712, the angleestimation logic 714, or both, are configured to receive an interruptservice routine (ISR) input 730. The ISR input 730 is a Boolean valueindicating an interrupt service routine mode. The demodulation logic712, the angle estimation logic 714, or both, stop generating outputsresponsive to receiving the ISR input 730.

FIG. 8 is a logic diagram 800 that illustrates an example of logic 802for voltage conditioning. The logic 802 is configured to receive voltagevalues from a sensor, condition the voltage values to remove hardwarebias, and output demodulated output values. In the particular exampleillustrated in FIG. 8, the logic 802 receives a voltage value measuredby the differential voltage sensor 316 corresponding to the sine coils424 of the fine resolver 344 and calculates the demodulated output 362for the fine resolver 344.

The logic 802 includes recursive midvalue analysis (RMVA) logic 812, avoltage filter 818, and a combiner 816. The RMVA logic 812 is configuredto determine a median (midvalue) of the last n number of input values orsamples, where n is an integer of greater than 1. The RMVA logic 812 isdescribed further with reference to FIG. 9.

During operation, the logic 802 receives multiple voltage values fromthe ADC 320 corresponding to the voltage at the sine coils 424 of thefine resolver 344 (e.g., a 16 speed resolver) measured by thedifferential voltage sensor 316. The RMVA logic 812 receives themultiple voltage values and outputs a midvalue voltage value 814 to thecombiner 816. The voltage filter 818, such as a low pass filter,generates a voltage bias output 820, sin tare as illustrated in FIG. 8,based on the midvalue voltage value 814 and the counter input 852. Thevoltage bias output 820 corresponds to a voltage bias of or imparted bythe hardware of the inertial measurement unit 102. For example, thevoltage bias output 820 may correct for a voltage drop associated withan FPGA of the inertial measurement unit 102.

In a particular implementation, the midvalue voltage value 814 isconverted to a higher bit value, such as from a 16 bit value to a 32 bitvalue. Converting from 16 bit to 32 bit (increasing a number of bits)reduces or eliminates a loss of precision caused by the voltage filter818. For example, any right shifts by the voltage filter 818 do notcause a loss of precision.

The combiner 816 generates an adjusted voltage value 822 (e.g., theconditioned voltage value 360 of FIG. 3) based on the midvalue voltagevalue 814 and the voltage bias output 820. As illustrated in FIG. 8, theadjusted voltage value 822 is a difference of the midvalue voltage value814 and the voltage bias output 820. The adjusted voltage value 822 maybe converted to a signed integer and shifted before demodulation. Theadjusted voltage value 822 may be output to the demodulation logic 826and stored as an input voltage value 824.

The demodulation logic 826 generates the demodulated output 362 based onthe input voltage value 824, the counter input 852, the width input 854,and the center input 856. Generation of the demodulated output 362 andexamples of the RMVA logic 812 demodulation logic 826 are describedfurther with reference to FIGS. 9-17.

FIG. 9 is a logic diagram 900 that illustrates an example of RMVA logic812 for recursive median value analysis of inputs, such as the ADCoutputs 358 of FIGS. 7 and 8. The RMVA logic 812 includes sliding windowlogic 904 configured to receive inputs and to store a last n number ofinputs. The sliding window logic 904 may include one or more registers(e.g., memories, caches, buffers, etc.) to store the last n number ofinputs. The sliding window logic 904 may perform a zero-order hold (ZOH)to store the last n number of inputs. The zero-order hold (ZOH) holdseach sample value for one sample interval. As new samples are received,each stored sample moves back one place and a last stored sample ispushed out. In the example illustrated in FIG. 9, the sliding windowlogic 904 stores the last seven inputs (p0-p6).

The RMVA logic 812 includes cascaded midvalue logic 906 configured todetermine a midvalue of the last n number of inputs, the midvaluevoltage value 814 of FIG. 8. The cascaded midvalue logic 906 includesmultiple logic blocks 912-916 each configured determine a midvalue for asubset of the last n number of inputs. An output of a first logic block912 is used as an input of the second logic block 914, and an output ofthe second logic block 914 is used an input for a third logic block 916,and so on. An output of a final logic block (i.e., the third logic block916 in the implementation illustrated in FIG. 9) is the median value forthe last n number of samples. FIG. 9 illustrates a particular example ofthe RMVA logic 812 where the multiple logic blocks 912-916 eachdetermine a midvalue of three inputs and three logic blocks 912-916 areused. In other implementations, more than three logic blocks or lessthan three logic blocks may be used and/or each logic block may processa different number of inputs from another logic block. Additionally oralternatively, each of the logic blocks 912 and 914 can determine amidvalue of 3 samples and the logic block 916 can determine a midvalueof the two outputs and a single sample.

FIG. 9 further illustrates an example of logic 922 for a 3 valuemidvalue determination, such as the first logic block 912. The logic 922allows parallel processing and faster determination of the median valuefor larger values of n. The first logic block 912 receives three inputscorresponding to a combination of samples and midvalue outputs. In thefirst logic block 912 illustrated in FIG. 9, the first logic blockreceives samples five, six, and seven (p4-p6) at inputs 1, 2, and 3respectively. The samples are compared to each other by a Booleancondition, illustrated as a greater than condition in FIG. 9. Threecomparison are used because three samples are used in the illustratedexample. These comparisons produce Boolean outputs that are used tocontrol switch outputs. To illustrate, the logic 922 compares the firstinput ([x]) to the third input ([z]) to generate a first Booleancondition ([xgz]) indicating whether or not the first input ([x]) isgreater than the third input.

The logic 922 uses a series of switches (h-h5) to determine the midvalueoutput. Each switch (h-h5) receives two input sample values and thecorresponding Boolean condition. For example, a first switch (h2)receives the first input ([x]) and the third input ([z]) and thecorresponding Boolean condition generated by comparing the first andthird inputs, i.e., the first condition ([xgz]). The first switch (h2)outputs the third input ([z]) based on the first condition indicatingfalse, i.e., x is not greater than z. A second switch (h4) receives thesecond input ([y]), the output (the third input ([z])) of the firstswitch (h2), and a Boolean condition that corresponds to the secondinput (e.g., the second Boolean condition or the third Booleancondition). The second switch (h4) compares the second input to thethird input based on one of the Boolean conditions that corresponds tothe second input, i.e., the second condition ([ygz]) in FIG. 9. Thesecond switch (h4) outputs the second input ([y]) to a fifth switch((h3) e.g., an output switch) based on the first condition indicatingtrue, i.e., y is greater than z.

Operation of a third switch (h) and a fourth switch (h5) mirroroperation of the first switch (h2) and the second switch (h4). Forexample, the third switch (h) receive the same inputs and condition inopposite order and accordingly, the third switch (h) outputs theopposite output value based on the first condition to the fourth switch(h5). To illustrate, the third switch (h) will output the first output([x]) if the first switch (h2) outputs the third output ([z]). One ofthe second switch (h4) and the fourth switch (h5) will always output thesecond input ([y]).

The fifth switch (h3) receives the output of the second and fourthswitches (h4 and h5) and compares them based on the other Booleancondition that corresponds to the second input ([y]). The fifth switch(h3) outputs the midvalue based on the second input ([y]), when thefirst input ([x]) is greater than the second input ([y]), the topswitches (h2 and h4) will output the midvalue. When the second input([y]) is greater than the first input ([x]), the bottom switches (h andh5) output will be the midvalue (given that a Boolean condition of trueor high selects the top input at each switch). As compared to sorting(e.g., by a sort function) the last n number of inputs, the RMVA logic812 and the logic 922 reduce firmware, circuit area, and processingspeed.

FIG. 10 is a logic diagram 1000 that illustrates an example of thedemodulation logic 826 of FIG. 8. The demodulation logic 826 isconfigured to generate the demodulated output 362 based on the inputvoltage value 824 of FIG. 8 (included in a signal 1012), the counterinput 852, the width input 854, and the center input 856. Thedemodulation logic 826 includes an accumulator logic 1014, mask logic1018, and output logic 1026.

The accumulator logic 1014 is configured to output an accumulated output1024 based on the input voltage values 824. For example, the accumulatorlogic 1014 is configured to accumulate values of the input voltagevalues 824 and output an accumulated output 1024 as an accumulation ofthe input voltage values 824. To illustrate, accumulation includesadding a current voltage value 824 to a sum of previous voltage values824. Additionally, the accumulator logic 1014 is configured to determineand adjust a sign of the accumulated output 1024 based on a flip input1022.

In some implementations, the accumulator logic 1014 adjusts and filtersmultiple input voltage values 824 to determine a particular accumulatedoutput 1024. For example, the accumulator logic 1014 is configured todetermine which input voltage values 824 of multiple input voltagevalues 824 to consider (or filter) based on a count enabled input 1020.Selected input voltage value 824 are accumulated and filtered inputvoltage values 824 are discarded (not accumulated). Additionally, theaccumulator logic 1014 is further configured to be reset responsive toreceiving a reset input 1016. Details of the accumulator logic 1014 aredescribed in further detail with respect to FIG. 11.

The mask logic 1018 is configured to generate the count enable input1020 based on the counter input 852, the width input 854, and the centerinput 856. Details of the mask logic 1018 are described in furtherdetail with respect to FIG. 14.

The output logic 1026 is configured to generate the demodulated output362 based on the accumulated output 1024 and the counter input 852. Forexample, the output logic 1026 is configured to determine thedemodulated output 362 as a midvalue of the accumulated outputs 1024based on to the counter input 852. Details of the output logic 1026 aredescribed in further detail with respect to FIG. 16.

During operation, the mask logic 1018 generates a take data output(i.e., a count enabled input 1020) based on the counter input 852, thewidth input 854, and the center input 856. Generation of the mask takedata output (the count enabled input 1020) and an example of mask logic1018 is described further with reference to FIGS. 14 and 15.

The accumulator logic 1014 receives the signal 1012 including multipleinput voltage values 824. The accumulator logic 1014 also receives thereset input 1016, the mask take data output (the count enabled input1020), and the flip input 1022. The accumulator logic 1014 generates theaccumulated output 1024 based on the multiple input voltage values 824,the reset input 1016, the mask take data output (the count enabled input1020), and the flip input 1022. The accumulator logic 1014 rectifies themultiple input voltage values 824 based on the flip input 1022 togenerate rectified values. The rectified values are output asaccumulated outputs 1024 based on the reset input 1016 and the mask takedata output (the count enabled input 1020). Generation of theaccumulated output 1024 and an example of accumulator logic 1014 isdescribed further with reference to FIGS. 11 and 12.

The flip input 1022 is a Boolean value indicating when to change a signof a square wave used in rectification. The flip input 1022 isconfigured to change the sign of the square wave used to rectify themultiple input voltage values 824. The flip input 1022 tracks thebehavior of the sine wave of the excitation signal of the resolver, suchas one of the excitation signals 352, 452 of FIGS. 3 and 4. For example,the sign of the square wave is positive when the counter input 852 valueis less than half of the counter period (illustrated as 2{circumflexover ( )}7−1 in FIG. 10), because a sine wave is positive for the firsthalf of the sine wave (i.e., 0 to 180 degrees). The sign of the squarewave is negative when the counter input 852 value is greater than halfof the counter period, because a sine wave is negative for the secondhalf of the sine wave (i.e., 181 to 360 degrees).

FIG. 11 is a logic diagram 1100 that illustrates an example of logic1102 for rectification with phase preservation. The logic 1102 includesflip generation logic 1104 and the accumulator logic 1014 of FIG. 10.The flip generation logic 1104 is configured to generate the flip input1022 as described with reference to FIG. 10.

The accumulator logic 1014 is configured to rectify the multiple inputvoltage values 824 based on the flip input 1022 to generate rectifiedvalues. The accumulator logic 1014 is configured to accumulate therectified values to generate the accumulated outputs 1024. Asillustrated in FIG. 11, the accumulator logic 1014 includes an inverter1112, multiple switches 1114, 1118, and 1120, and a summer 1116.

The inverter 1112 is configured to invert the multiple input voltagevalues 824. For example, the inverter 1112 is configured to multiply themultiple input voltage values 824 by −1. The multiple switches 1114,1118, and 1120 are configured to output values based on control inputs,i.e., the flip input 1022, the count enabled input 1020, and the resetinput 1016 respectively. The summer 1116 is configured to add an outputof the first switch 1114 and a delayed output of the third switch 1120.A unit delay 1122 is configured to delay the output of the third switch1120 for one sample. For example, the unit delay 1122 performs azero-order hold.

During operation, the flip generation logic 1104 generates the flipinput 1022 based on comparing the counter input 852 to a counter midvalue (e.g., 2{circumflex over ( )}7−1 for an 8 bit counter 842). Whenthe counter input 852 is greater than the counter mid value, the flipgeneration logic 1104 generates the flip input 1022 indicating true(e.g., 1) or to flip the sign of the input voltage values 824.

The first switch 1114 receives the input voltage values 824 of thesignal 1012, inverted input voltage values 824 from the inverter 1112,and the flip input 1022. The first switch 1114 outputs the invertedinput voltage values 824 based on the flip input indicating 1, whichcorresponds to a second half of a sine wave of the excitation signalwhere the amplitude of the sine wave is negative. Thus, the flipgeneration logic 1104, the inverter 1112, and the first switch 1114 actto rectify the input voltage values 824 and preserve a phase of theexcitation signal. The flip generation logic 1104, the inverter 1112,and the first switch 1114 function to multiply the input voltage values824 by a square wave of [1,−1] that is in phase with the excitationsignal.

The summer 1116 receives the rectified input voltage values 1152 fromthe first switch 1114 and combines the rectified input voltage values1152 with a delayed output 1158 of the unit delay 1122 to generate acombined output 1154. The combined output 1154 is provided to the secondswitch 1118. The second switch 1118 outputs the combined output 1154 orthe delayed output 1158 based on the count enabled input 1020. When thecount enabled input 1020 indicates true or to take data, the secondswitch 1118 outputs the combined output 1154, as shown in FIG. 11.Alternatively, the second switch 1118 outputs the delayed output 1158when the count enabled input 1020 indicates false or to not take data.

The output of the second switch 1118 is provided to the third switch1120. The third switch 1120 outputs a null value 1124 (e.g., 0) when thereset input 1016 indicates true or to reset the accumulator logic 1014.The third switch 1120 outputs the output of the second switch 1118 as anoutput 1156 of the third switch when the reset input 1016 indicatesfalse or not to reset the accumulator logic 1014. The output 1156 of thethird switch 1120 is provided to the unit delay 1122. The unit delay1122 delays the output 1156 of the third switch 1120 and provides thedelayed output 1158 to the summer 1116 and the second switch 1118. Thedelayed output 1158 is also output as the accumulated output 1024.

Providing the delayed output 1158 to the summer 1116 and the secondswitch 1118 outputting the combined output 1154 responsive to the countenabled input 1020 indicating true, function to increase the delayedoutput 1158 iteratively when new input voltage values 824 are received.Thus, when the delayed output 1158 is output as the accumulated output1024 (i.e., before the accumulator logic 1014 is reset), the accumulatedoutput 1024 represents an accumulation of input voltage values 824 thatcorrespond to periods of time when the mask logic 1018 outputs the countenabled input 1020 indicating to take or accumulate data. As illustratedin FIG. 10 and further described with reference to FIGS. 16 and 17, theoutput logic 1026 generates the demodulated outputs 362 based on theaccumulated outputs 1024 and outputs the demodulated outputs 362 basedthe counter input 852.

Although many inputs have been described as Boolean inputs to controllogic gates or switches, the logic gates or switches may be implementedas transistors and receive logical high and low signals (e.g., high andlow voltages) representing the Boolean inputs.

FIG. 12 illustrates a diagram 1200 of exemplary signals generated duringdemodulation. Diagram 1200 includes three graphs 1202-1206. A firstgraph 1202 illustrates an ADC output signal 1212 (demodulator inputs)corresponding to sine wave outputs of a resolver, such as the fineresolver 344 of FIG. 3. The ADC output signal 1212 includes the ADCoutputs 358 (or the conditioned voltage values 360) of FIG. 3. A secondgraph 1204 illustrates a rectified ADC output signal 1214 correspondingto the first graph 1202. In the second graph, the rectified ADC outputsignal 1214 has the same phase as the ADC output signal 1212. The ADCoutput signal 1212 is rectified (e.g., multiplied) by a square wave,instead of the excitation signal (e.g., a sine wave), to preserve thephase of the ADC output signal 1212 during rectification and to increaseprecision. The rectified ADC output signal 1212 includes or correspondsto the rectified input voltage values 1152 of FIG. 11, such as multiplerectified ADC outputs. A third graph 1206 illustrates a demodulationoutput signal 1216 of demodulated sine amplitude values. The demodulatedsine amplitude values (including the sign information, i.e., positive ornegative) are used to calculate the position of the drive shaft of themotor and correspond to the demodulated outputs 362 of FIG. 3. Asillustrated in the third graph, the demodulation output signal 1216 hasa delay (is out of phase) from the ADC output signal 1212 and therectified ADC output signal 1214. The delay is shown and explained ingreater detail with respect to FIG. 13.

Rectification with a square wave may introduce error in an area 1222where the amplitudes of the excitation signal changes from positive tonegative values (i.e., crosses zero) and the amplitudes of the ADCoutput signal 1212 approach zero. Accumulation of voltage values,masking voltage values, outputting median accumulated values, or acombination thereof reduces or eliminates this error or drawback, asshown by the demodulation output signal 1216 not including error in anarea corresponding to the area 1222.

FIG. 13 illustrates exemplary graphs 1302 and 1304 of signals of thecoarse resolver 342 of FIG. 3. A first graph 1302 includesrepresentations of sine and cosine values output by the coarse resolver342. The first graph 1302 may represent the sine and cosine values 1312,1314 sampled by the ADCs 318, 320 from differential voltage outputs ofthe coarse resolver 342 of FIG. 3. Prior to a first time T1, the coarseresolver 342 is not moving. After the first time T1, the coarse resolver342 begins moving and causes changes in the sine and cosine values 1312,1314.

A second graph 1304 includes representations of demodulated sine andcosine values 1322, 1324 corresponding to the sine and cosine values1312, 1314 of the first graph 1302. Demodulation of the sine and cosinevalues 1312, 1314 to generate the demodulated sine and cosine values1322, 1324 involves removing or deemphasizing the excitation signal,such as one of the excitation signals 352, 452 of FIGS. 3 and 4. Thesine and cosine values 1312, 1314 correspond to the ADC outputs 358 (orthe conditioned voltage values 360) of FIG. 3 associated with the coarseresolver 342. The demodulated sine and cosine values 1322, 1324correspond to the demodulated outputs 362 of FIG. 3 associated with thecoarse resolver 342. As compared to the signals 1212-1216 of FIG. 12,the values 1312, 1314, 1322, and 1324 correspond to the coarse resolver342 (as opposed to the signals 1212-1216 which correspond to the fineresolver 344).

As illustrated in the second graph 1304, changes in the sine and cosinevalues 1312, 1314 are reflected two cycles later in the demodulated sineand cosine values 1322, 1324, i.e., the changes are delayed by twocycles. It takes one cycle (e.g., from peak amplitude to peak amplitudeof sine or cosine wave) to measure the sine and cosine values 1312, 1314and a second cycle to process a midvalue of the measured values duringdemodulation. To illustrate, a change in value (due to movement of thecoarse resolver 342) of the sine and cosine values 1312, 1314 occurringduring a first cycle from a first time T1 to a second time T2, is notreflected in the demodulated sine and cosine values 1322, 1324 until athird cycle from a third time T3 to a fourth time T4. Accordingly, thedemodulation logic 826 imparts a relatively small delay (two cycles) andenables a relatively large increase in precision (multiple orders ofmagnitude) as compared to conventional demodulation systems, furtherdescribed with respect to FIGS. 20-25.

FIG. 14 is a logic diagram 1400 that illustrates an example of the masklogic 1018 of FIG. 10. The mask logic 1018 is configured to generate thecount enable input 1020 based on the counter input 852, the width input854, and the center input 856. As illustrated in the example of FIG. 14,the mask logic 1018 includes multiple combiners 1410-1418, comparisonconditions 1422-1428, and logic gates 1432-1436.

During operation, the mask logic 1018 receives the width input 854, thecenter input 856, and the counter input 852. The mask logic 1018 alsoreceives or determines an offset input 1402. The offset input 1402 maybe determined based on a number of bits of the inputs 852-856. Forexample, when the inputs 852-856 correspond to 8 bit values, a value ofthe offset input 1402 is 128, i.e., 2{circumflex over ( )}(8−1). In aparticular implementation, the inputs 852-856 are 8 bit unsigned integervalues. The difference between the center input 856 and the width input854 represents half of the width input 854.

The mask logic 1018 includes two logic chains 1404, 1406 configured togenerate Boolean outputs. The mask logic 1018 generates an output basedon a logical operation of the Boolean outputs from the two logic chains1404, 1406. As illustrated in FIG. 14, the mask logic 1018 generates atake data Boolean output based on an “OR” logic operation of the Booleanoutputs from the two logic chains 1404, 1406. In other implementations,other logical operations, such as NOR, AND, etc., may be used. A valueof the take data Boolean output (i.e., the count enable input 1020)indicates to the accumulator logic 1014 to take data or not take data,as described with reference to FIG. 10.

During operation, the mask logic 1018 receives the three inputs 852-856.A value of the counter input 852 may change each time a counter value ofthe counter 842 is increased, such as based on a clock pulse. Thecounter input 852 is reset based on a reset pulse or reaching a maximumcounter value. The width and center inputs 854, 856 may be constant andinput by a user or fixed before operation. The offset input 1402 may beinput by a user, fixed before operation, or determined based on one ormore of the inputs 852-856.

A first combiner 1410 of the first logic chain 1404 generates adifference between the center input 856 and half of the width input 854(i.e., a [width]/2 input indicating a deviation or delta from the centerinput 856 that defines a data window 1514, as illustrated in FIG. 15). Afirst comparator 1422 compares the counter input 852 to the differencebetween the center input 856 and half of the width input 854 based on afirst comparison condition. A second combiner 1412 generates a sum ofthe center input 856 and half of the width input 854. A secondcomparator 1424 compares the counter input 852 is compared to the sum ofthe center input 856 and half of the width input 854 based a secondcomparison condition. As illustrated in FIG. 14, the first comparisoncondition is a greater than or equal to condition and the secondcomparison condition is a less than or equal to condition. In otherimplementations, other comparison conditions may be used. Outputs of thecomparators 1422, 1424 are input to a first logic gate 1432. The firstlogic gate 1432 outputs a first intermediary output to a third logicgate 1436 (the output logic gate) based on a logic rule of the firstlogic gate 1432.

A third combiner 1414 of the second logic chain 1406 generates a sum ofthe center input 856 and the offset input 1402. A fourth combiner 1416generates a difference between, the sum of the center input 856 and theoffset input 1402, and half of the width input 854. A third comparator1426 compares the counter input 852 to the difference between the sumand half of the width input 854 based a third comparison condition. Asillustrated in FIG. 14, the third comparison condition is the same asthe first comparison condition. A fifth combiner 1418 generates a sum ofthe center input 856, the offset input 1402, and half of the width input854. A fourth comparator 1428 compares the counter input 852 to the sumof the inputs 854, 856, and 1402 (i.e., the sum of the center input 856and the offset input 1402 plus half of the width input 854) based afourth comparison condition. As illustrated in FIG. 14, the fourthcomparison condition is the same as the second comparison condition.Outputs of the comparators 1426, 1428 are input to a second logic gate1434. The second logic gate 1434 outputs a second intermediary output tothe third logic gate 1436 (output logic gate) based on a logic rule ofthe second logic gate 1434.

The third logic gate 1436 generates the take data output (the countenabled input 1020) based on a logic rule of the third logic gate 1436and provides the take data output (the count enabled input 1020) to theaccumulator logic 1014 of FIG. 10. As illustrated in FIG. 14, the logicrules of the first and second logic gates include AND logic and thelogic rule of the third logic gate 1436 includes OR logic. In otherimplementations, other logic rules, additional logic gates, or both, maybe used to regenerate the take data output (the count enabled input1020).

FIG. 15 is a diagram 1500 that illustrates an example of masked data andaccumulated data for demodulation. Diagram 1500 illustrates an examplegraph depicting accumulator signal 1522 (representing outputs 1156 ofthe third switch 1120), and a demodulated output signal 1524(representing the accumulated outputs 1024) of the accumulator logic1014, and masked regions 1512. Diagram 1500 also illustrates the widthinput 854 and the center input 856 in data windows 1514 between twoparticular masked regions 1512. In the diagram 1500, the center input856 is positioned in a first data window 1514 of a cycle correspondingto (generated by outputs of) the first logic chain 1404 of FIG. 14, andthe width and offset inputs 854, 1402 are position in a second datawindow 1514 of the cycle corresponding to (generated by outputs of) thesecond logic chain 1406 of FIG. 14. The cycle corresponds to a period ofthe sine wave and the counter input 852 being reset.

The accumulator signal 1522 increases when the count enabled input 1020indicates to take or accumulate data filtered data, i.e., in the datawindows 1514 between masked regions 1512. The accumulator signal 1522does not increase during masked regions 1512 when the count enabledinput 1020 indicates to refrain from taking or accumulating data. Theaccumulator logic 1014 outputs the demodulated output signal 1524(values thereof) during the masked regions 1512 (i.e., every secondmasked region 1512) after stopping taking data in the masked regions1512. After outputting a value of the demodulated output signal 1524,the accumulator logic 1014 is reset (i.e., a value of the accumulatorsignal 1522 reverts to 0) during the masked regions 1512. Thus, in theexample illustrated in FIG. 15, the accumulator logic 1014 accumulatesdata during two data windows 1514 of a cycle and outputs the accumulatedoutputs 1024 once per cycle.

FIG. 16 is a logic diagram 1600 that illustrates an example of theoutput logic 1026 of FIG. 10. The output logic 1026 is configured tooutput a median value of the accumulated output 1024 values determinedby the accumulator logic 1014 as the demodulated output 362 based on theaccumulated output 1024 and the counter input 852. Operations of theaccumulator logic 1014 and the output logic 1026 may be performed by anaccumulator (e.g., accumulator circuitry).

As illustrated in the example of FIG. 16, the output logic 1026 includesa switch 1612 and the RMVA logic 812. As compared to the RMVA logic 812of FIGS. 8 and 9, the RMVA logic 812 is configured to output valuesresponsive to or based on a Boolean input value 1628. The RMVA logic 812outputs the demodulated output 362 responsive to the Boolean input value1628 indicating true.

The output logic 1026 is further configured to generate the Booleaninput value 1628 based on the counter input 852. As illustrated in FIG.16, a unit delay 1622 delays the counter input 852 by one sample tosynchronize timings with inputs to the RMVA logic 812. The Boolean inputvalue 1628 is generated based on a comparison of a delayed counter value1626 to a comparison condition 1624. As illustrating in FIG. 16, thecomparison condition 1624 is an equal to zero condition, and the Booleaninput value 1628 indicates true when the delayed counter value 1626satisfies the comparison condition 1624 (i.e., is equal to zero). TheBoolean input value 1628 indicates false when the delayed counter value1626 does not satisfy the comparison condition 1624 (i.e., not equal tozero).

During operation, the output logic 1026 receives the counter input 852.The counter input 852 is provided to the switch 1612, and the switch1612 outputs the accumulated output 1024 or a delayed accumulator output1654 based on the counter input 852. As illustrated in FIG. 16, theswitch 1612 outputs the delayed accumulator output 1654 based on thecounter input 852 satisfying a condition of not equal to zero. An output1652 of the switch 1612 is provided to a unit delay 1614. The unit delay1614 holds the output 1652 for a time period corresponding to one ormore samples to generate the delayed accumulator output 1654. Thedelayed accumulator output 1654 is provided to the switch 1612 and theRMVA logic 812.

When the counter input 852 is equal to zero, the value of theaccumulated output 1024 is reset to zero, as explained with reference toFIG. 11. While the counter input 852 is not equal to zero, the outputlogic 1026 stores the delayed accumulator output 1654 value until theaccumulated output 1024 is received by the delay unit 1614 again i.e.,when the counter input 852 is equal to 0. Thus, the unit delay 1614(e.g., a register or shadow register) holds the value of the delayedaccumulator output 1654 for one cycle, i.e., until a new accumulatedoutput 1024 is received at the start of the next resolver excitationsignal cycle, i.e., when the counter input 852 is equal to 0. The valueof the delayed accumulator output 1654 corresponds to the previous finalaccumulator value of the accumulated output 1024 before the accumulatedoutput 1024 was reset to zero. Because the RMVA logic 812 takes thevalue of the delayed accumulator output 1654 while the delayedaccumulator output 1654 is being held by the switch 1612 and the delayunit 1614 (i.e., sampled by the RMVA logic 812 one sample after thecounter input 852 is zero), the RMVA logic 812 receives the previousfinal accumulator value of the accumulated output 1024 before it wasreset to zero.

The RMVA logic 812 determines a midvalue of the last n delayedaccumulator outputs 1654 and outputs the midvalue of the last n delayedaccumulator outputs 1654 as the demodulated output 362 responsive to theBoolean input value 1628 indicating true. In some implementations, theRMVA logic 812 of FIG. 8 uses the same number n for the last n inputs asthe RMVA logic 812 of FIG. 16. In other implementations, the RMVA logic812 of FIG. 8 uses a different number n for the last n inputs as theRMVA logic 812 of FIG. 16. In a particular implementation, the RMVAlogic 812 determines a median demodulated output 1656 based on the lastthree delayed accumulator outputs 1654.

FIG. 17 is a diagram 1700 that illustrates an example of accumulatoroutputs for the demodulation circuitry. The accumulator, such as theaccumulator logic 1014 and the output logic 1026 of FIG. 10, maygenerate multiple intermediary values and an output value. Asillustrated in FIG. 17, the accumulator generates the accumulator signal1522 (by accumulating values corresponding to outputs 1156 of the thirdswitch 1120), generates the demodulated output signal 1524(corresponding to accumulated outputs 1024), and generates thedemodulated sine outputs 362 (corresponding median values of thedemodulated output signal 1524 or the median demodulated output 1656),as described with reference to FIGS. 10-16. The demodulated outputsignal 1524 and the demodulated sine outputs 362 include signinformation (i.e., positive or negative).

FIG. 18 is a logic diagram 1800 that illustrates an example of logic1802 for combining resolver outputs of the dual speed resolver 312. Thelogic 1802 is configured to combine outputs from each resolver 342, 344of the dual speed resolver 312 to determine an initial position in thefirst domain or an absolute initial position and to determine subsequentpositions based on the fine resolver alone. Determining subsequentpositions (e.g., changes in position from the initial position) based onthe fine resolver alone increases precision over dual speed resolversthat determine subsequent positions based on outputs from the coarseresolver. Operations of the logic 1802 may be performed by the anglecombination circuitry 542 of FIG. 5.

The logic 1802 receives the angle estimates 364 in a second domain basedon fine position signals from the fine resolver 344, referred to assecond domain angle estimates 1812 in FIG. 18. The second domain angleestimates 1812 received from the fine resolver 344 are in the seconddomain (e.g., a 16 speed domain) associated with the fine resolver 344.The logic 1802 differentiates the second domain angle estimates 1812 todetermine a change in position (speed) in the second domain. Forexample, a previous position value 1814 is subtracted from a currentposition value 1816 of the second domain angle estimates 1812 togenerate a change in position value 1818. The change in position value1818 may be converted to a 32 bit signed integer and delayed to accountfor a processing pipeline or flow of the circuitry used to perform thelogic 1802. The change in position value 1818 is right shifted by 4(divided by 16) to convert the change in position value 1818 into thefirst domain to generate a converted change in position value 1820(e.g., transformed fine position signals of the fine resolver 344 ofFIG. 3). The converted change in position value 1820 represents a changein position determined by the fine resolver 344 with respect to thefirst domain.

The logic 1802 receives an estimated initial position 1822 determinedbased on coarse position signals of the coarse resolver 342. Theestimated initial position 1822 corresponds to a first or initial angleestimate of the angle estimates 364 of the coarse resolver 342. Theestimated initial position 1822 represents the initial positiondetermined by the coarse resolver 342 and is in the first domain. Aswitch 1826 (e.g., an initialization switch) is configured to output theestimated initial position 1822 when a Boolean input 1824 (init_rslv)indicates true, i.e., the dual speed resolver 312 is in aninitialization mode. The switch 1826 is configured to output positionfeedback (e.g., a previously determined position, such as a delayedcombined estimated angle 1844) when the Boolean input 1824 (init_rslv)indicates false, i.e., the dual speed resolver 312 is not in theinitialization mode.

The converted change in position value 1820 and the estimated initialposition 1822 are combined by a combiner 1830 (e.g., adder or otherarithmetic circuitry) to generate a combined estimated angle 1832 (e.g.,an estimated position of the motor). The combined estimated angle 1832includes or corresponds to the estimated position 366 of FIG. 3. In someimplementations, the combiner 1830 further receives a drift correctionvalue 1828 and is configured to generate the combined estimated angle1832 based further on the drift correction value 1828. In suchimplementations, the combined estimated angle 1832 is delayed, such asby a zero order hold, to generate the delayed combined estimated angle1844. The delayed combined estimated angle 1844 is provided to the driftcorrection circuitry 544 of FIG. 5 to generate the drift correctionvalue 1828. Generation of the drift correction value 1828 is describedwith reference to FIG. 19. For a first or initial combined estimatedangle 1832 (i.e., an initial position), the drift correction value 1828is zero because the drift correction circuitry 544 has not yet receiveda combined estimated angle 1832 and begun to provide the driftcorrection value 1828.

The combined estimated angle 1832 is provided as an output of the logic1802. In some implementations, the logic 1802 generates one or moreadditional outputs based on the combined estimated angle 1832. Forexample, the combined estimated angle 1832 is combined with a rotoroffset 1852 for commutation of the rotor. To illustrate, the rotoroffset 1852 for commutation of the motor 124 accounts for themisalignment between the resolver and rotor magnets of the motor 124.Additionally or alternatively, the combined estimated angle 1832 iscombined with a rotor offset 1862 for servo control of the rotor, suchas a rotor servo offset. To illustrate, the rotor offset 1862 for servocontrol of the motor 124 accounts for an offset of a state in which thegimbal (e.g., one of the gimbals 122 of FIG. 1) is pointing upwardsrelative to a horizon or opposite a gravitational force. In a particularimplementation, the rotor offsets 1852, 1862 are a constant values andare input by a user.

These rotor offsets 1852, 1862 are commonly referred to as taring theresolver outputs. Taring the resolver outputs helps improve precisionand precision taring is often needed in conventional dual resolvers.However, taring has reduced effects (reduced improvements) and lessprecise taring may be used when reducing noise due to time coordinationof excitation signals and current drive switching, when dither is addedto the excitation signal, or both, as described with reference to FIGS.22-28. Tared outputs 1856, 1866 may be provided to other components toadjust for commutation and servo control of the rotor. For example, theservo offset output 1866 may be provided to the feedback control system232 of FIG. 2 for initialization of the inertial measurement unit 102,as described with reference to FIG. 32. As another example, the taredoutputs 1856, 1866 may indicate the estimated position of the motor 124.To illustrate, when the combined estimated angle 1832 indicates theposition of the dual speed resolver 312 of FIG. 3, the combinedestimated angle 1832 may be adjusted to indicate the position of themotor 124. The combined estimated angle 1832 is adjusted/tared toindicate the estimated position of the motor or components thereof, suchas the mechanical or magnetic rotor position of FIG. 3. Additionally oralternatively, the tared outputs 1856, 1866 may be used to determine theestimated RPM 368 of the motor 124.

The logic 1802 generates subsequent position outputs (outputs after aninitial position is determined or after initialization process)independent of subsequent coarse resolver 342 position inputs. Toillustrate, when the Boolean input 1824 indicates that initializationprocess is completed (e.g., when the Boolean input 1824 is equal tozero), the switch 1826 “filters out” the subsequent coarse resolver 342position inputs (i.e., the estimated initial position 1822) and providesthe previously calculated angle (the delayed combined estimated angle1844). Said another way, the logic 1802 determines subsequent changes inposition based on the subsequent fine resolver 344 outputs of the fineresolver 344 alone.

FIG. 19 is a logic diagram 1900 that illustrates an example of logic1902 for a drift corrector of the dual speed resolver 312. Operations ofthe logic 1902 may be performed by the drift correction circuitry 544 ofFIG. 5. The logic 1902 is configured to generate the drift correctionvalue 1828 based on the delayed combined estimated angle 1844, thesecond domain angle estimates 1812 and a 180 degree bias value 1918.

The logic 1902 receives the angle estimates 364 of the fine resolver 344of the dual speed resolver 312 as the second domain angle estimates1812. The logic 1902 further receives the delayed combined estimatedangle 1844 (e.g., a delayed version of the combined estimated angle 1832generated by the logic 1802 of FIG. 18). The delayed combined estimatedangle 1844 is in the first domain (e.g., a 1 speed domain) associatedwith the motor 124 and the coarse resolver 342. The logic 1902 convertsthe delayed combined estimated angle 1844 to the second domain (e.g., a16 speed domain) associated with the fine resolver 344 of the dual speedresolver 312. As illustrated in FIG. 9, the delayed combined estimatedangle 1844 is left shifted by 4 (multiplied by 16) to generate aconverted estimated angle 1914 (e.g., transformed initial positionoutputs).

The converted estimated angle 1914 and the 180 degree bias value 1918are subtracted from the second domain angle estimates 1812 to generatean error value 1920. In the implementation illustrated in FIG. 9, thelogic 1902 uses 32 bit integers and the 180 degree bias value 1918corresponds to a value of 2{circumflex over ( )}31−1 (or 2{circumflexover ( )}(number of bits−1)−1). The 180 degree bias value 1918 providesprotection against noise and integer math errors which cause incorrectalignment of the resolvers 342, 344 during initialization. The noise andinteger math errors may be caused by differentiation of the seconddomain angle estimates 1812.

The error value 1920 may be converted into a signed integer, such as a32 bit signed integer. Proportional gain 1922 and integral gain 1924 areapplied to the error value 1920 by proportional gain circuitry andintegral gain circuitry. In some implementations, the proportional gain1922 and integral gain 1924 are generated based on the error value 1920.As illustrated in FIG. 19, the proportional gain 1922 is generated byright shifting the error value 1920 by 15 (dividing by 32768), and theintegral gain 1924 is generated by a switch 1926. The switch 1926outputs the integral gain 1924 having a value of 1 or −1 based oncomparing the error value 1920 to a comparison condition of greater thanor equal to zero.

The proportional gain 1922 and the integral gain 1924 are combined(added) by a combiner 1928 to generate a drift correction value 1828.The drift correction value 1828 may be stored, provided to the logic1802, or both. For example, the drift correction value 1828 is providedto the combiner 1830 of the logic 1802 to determine a next combinedestimated angle 1832, as described with reference to FIG. 18. The nextcombined estimated angle 1832 adjusts or corrects a subsequent combinedestimated angle 1832 (combined estimated angle 1832 for subsequentpositions) to account for the lower precision of the coarse resolveroutputs used in generating the combined estimated angle 1832 for theinitial position. Accordingly, the logic 1902 may correct or reduceerrors that occur when combining resolvers of different speeds in a dualspeed resolver, such as the dual speed resolver 312 of FIG. 3.

FIG. 20 includes a diagram 2000 illustrating an estimated angle 2002 ofthe resolver system 204 and an actual angle 2004 of the motor. Theestimated angle 2002 may include or correspond to the estimated position366 of FIG. 3 or the combined estimated angle 1832 of FIG. 18.Alternatively, the estimated angle 2002 may include or correspond to anadjusted (tared) estimated position 366 (e.g., one of the tared outputs1856, 1858 of FIG. 18) or the combined estimated angle 1832 of FIG. 18.Diagram 2000 illustrates the estimated angle 2002 output by the resolversystem 204 during and after an initialization process or mode.

At a first time T1 of FIG. 20, the resolver system 204 is activated andbegins an initialization mode or process. During the initialization modeor process, the resolver system 204 generates an initial position output2012 of the estimated angle 2002 at a second time T2 of FIG. 20 based onthe coarse resolver outputs. After the second time T2 of FIG. 20, theresolver system 204 begins to utilize the drift correction circuitry 544to correct the estimated angle 2002. At a third time T3 of FIG. 20, thedrift correction circuitry 544 has adjusted for an initial error of thecoarse resolver 342 and is locked on to the position of the stationarymotor. After the third time T3, the resolver system 204 generates asubsequent position output 2014 of the estimated angle 2002. At a fourthtime T4 of FIG. 20, the motor begins to move. At a fifth time T5 of FIG.20, the motor stops moving.

FIG. 21 includes a diagram 2100 that depicts an enlarged view of thediagram 2000 of FIG. 20. Diagram 2100 better illustrates the delay andprecision of the resolver system 204. Diagram 2100 depicts a small errorat the third time T3 after the drift correction circuitry 544 hasadjusted for the initial error of the coarse resolver 342 and is lockedon to the position of the stationary motor. As described with referenceto FIG. 20, at the fourth time T4, the motor begins to move. However,the resolver system 204 does not detect the movement and output an angleestimate corresponding to the movement until a fifth time T5 of FIG. 21,because of the demodulation processing delays described with referenceto FIGS. 12 and 13. After the fifth time T5, the resolver system 204begins to track the movement of the motor that occurred at the fourthtime T4. Diagram 2100 illustrates a precision of 0.01 arc-sec.

FIG. 22A is a diagram 2202 that illustrates angles determined based onresolver outputs using an excitation signal without dither, such as theexcitation signal 452 of FIG. 4. FIG. 22A depicts activation of themotor at about 0.02 seconds and deactivation at about 0.0375 seconds. InFIG. 22A, the estimated angle 2002 of the drive shaft of the motortracks the actual angle 2004 of the drive shaft with a precision ofabout 1.23 arc-sec. The motor is reactivated and resumes moving at about0.06 seconds and the resolver system 204 continues to track the motor.

FIG. 22B is a diagram 2204 that illustrates ADC outputs 2252 generatedbased on resolver outputs generated by the excitation signal withoutdither. The ADC outputs 2252 are substantially symmetrical and uniformand the ADC outputs 2252 latch on to single bit for relatively longperiods of time. Latching onto a single bit for relatively long periodsof time prevents oversampling from increasing precision because itcauses the accumulated value to not be zero and/or to not change.

FIG. 23 is a diagram 2300 that illustrates an example of an excitationsignal including dither, such as the dithered excitation signal 352 ofFIG. 3. In FIG. 23, the dithered excitation signal 352 still repeatslike a regular sine wave, but the amplitude of the dithered excitationsignal 352 has local variations. For example, the amplitude of ditheredexcitation 352 has sinusoidal fluctuations over the course of a sinewave of the base excitation signal 452 (e.g., a first harmonic). Toillustrate, the amplitude increases at times during transitions frompeaks to valleys and decreases at times during transitions from valleysto peaks. As illustrated in FIG. 23, the excitation signal 452 isdithered based on a high order even harmonic of the excitation signal452 to create the dithered excitation signal 352. Thus, the ditheredexcitation signal 352 includes “miniature sine wave” deviations from theexcitation signal 452. Because a high order even harmonic (e.g., 16^(th)harmonic) is used, the dither does not affect or alter the meanamplitude of the sine wave because the base excitation signal 452 andeven harmonic have the same phase and because a harmonic signal has thesame function as the base excitation signal 452. Accordingly, the ditherdoes not introduce noise that results in additional errors and adecrease in precision.

FIG. 24 is a diagram 2400 that illustrates ADC outputs 2402 generatedbased on resolver outputs generated by a dithered excitation signal. Ascompared to the ADC outputs 2252 generated based on the excitationsignal without dither of FIG. 22B, the ADC outputs 2402 generated basedthe dithered excitation signal have more variability and the ADC outputs2402 do not latch on to the same value for as long or as much. Thus, theaccumulated output 1024 can fluctuate and be zero. Accordingly, theestimated angle 2002 can fluctuate from overestimating andunderestimating the actual angle 2004, as shown in FIG. 25.

FIG. 25 is a diagram 2500 that illustrates angles determined based onresolver outputs generated based on the dithered excitation signal 352of FIG. 23. In FIG. 25, the estimated angle 2002 of the drive shaft ofthe motor tracks the actual angle 2004 of the drive shaft. The ditheredexcitation signal 352 causes the estimated angle 2002 to switch betweenoverestimating and underestimating the actual angle 2004 of the driveshaft. Thus, by averaging the estimated angle 2002 (e.g., averaging overa 1 second interval), the dithered excitation signal 352 enables theresolver system 204 to provide 0.01 arc-sec of precision. A precision of0.01 arc-sec represents a multiple order of magnitude increase inprecision as compared to the precision of the estimated angle 2002 ofFIG. 22A.

FIG. 26 is a logic diagram 2600 that illustrates an example of logic2602 for excitation signal generation. The logic 2602 is configured togenerate excitation signals to be provided to an active sensor, such asthe dual speed resolver 312 of FIG. 3. For example, the logic 2602outputs digital differential outputs to the DAC 310, which converts thedigital differential outputs into the excitation signals provided to therotating primary coil 422 of the resolvers 342, 344. Operations of thelogic 2602 may be performed by the excitation signal generation system202 of FIG. 2 (e.g., excitation signal generation circuitry).

The logic 2602 enables generation of different types of excitationsignals, including the dithered excitation signal 352 of FIG. 3 or theexcitation signal 452 of FIG. 4. The logic 2602 includes one or morelogic chains configured to generate an excitation signal. As illustratedin FIG. 26, the logic 2602 includes three logic chains 2612-2616. Eachlogic chain 2612-2616 is directed to generating a portion of theexcitation signal. In other implementations, the logic 2602 may includeadditional logic chains or fewer logic chains. In the exampleillustrated in FIG. 26, the logic 2602 includes logic chains 2612-2616,two of which (logic chains 2612 and 2614) are active.

A first logic chain 2612 is configured to generate a first orderharmonic (i.e., the base excitation signal 452). A second logic chain2614 is configured to generate dither. As illustrated in FIG. 26, thesecond logic chain 2614 is configured to generate a high order evenharmonic of the base excitation signal. An example of a high order evenharmonic is a 16^(th) harmonic. Other high order even harmonics may beused depending on the sampling rate of the ADC's and DAC's, thefrequency of the hardware, or a combination thereof. For example, otherimplementations may utilize an 8^(th) harmonic, a 10^(th) harmonic, a12^(th) harmonic, an 18^(th) harmonic a 20^(th) harmonic, a 24^(th)harmonic, a 32^(nd) harmonic, a 64^(th) harmonic, etc., or a combinationthereof. A third logic chain 2616 is configured to generate a low orderodd harmonic of the base excitation signal. As illustrated in FIG. 26,the third logic chain 2616 is configured to generate a third harmonic ofthe base excitation signal 452. Addition of one or more low order oddharmonic tends to “square” the excitation signal. Other low order oddharmonics may be used, alone or in combination with the third harmonic,to square the excitation signal. Squaring the excitation signal canincrease the signal to noise ratio of sensor outputs.

During operation, the first logic chain 2612 generates the baseexcitation signal 452. As illustrated in FIG. 26, the first logic chain2612 generates the base excitation signal 452 as a differential signal(including positive and negative signal components) The first logicchain 2612 receives two inputs: a sine function at input 2652 and anamplitude setting for the sine function at input 2654. In someimplementations, the first logic chain 2612 reduces a number of bits ofthe sine function, the amplitude setting, or both. Additionally oralternatively, the first logic chain 2612 shifts the amplitude settingby addition of a user defined value, e.g., 0.1 in FIG. 26. The firstlogic chain 2612 multiplies the sine function by the shifted amplitudesetting to generate an intermediary sine wave.

The first logic chain 2612 generates differential outputs (e.g., apositive output and a negative output) based on the intermediary sinewave. For example, the first logic chain 2612 includes conversion logic2622 configured to generate the differential outputs based on generatinga positive value and a negative value for each value of the intermediarysine wave.

In some implementations, the conversion logic 2622 is further configuredto convert the positive and negative outputs to unsigned values and tofurther reduce a number of bits of the positive and negative outputs forprocessing by the DAC 310 of FIG. 3. Reducing a number of bits of thepositive and negative outputs does not lower the precision of inertialmeasurement unit 102 when the positive and negative outputs have morebits than the DAC 310 is configured to process. Additionally, the firstconversion logic 2622 adjusts the positive and negative outputs based onaddition of a mid-voltage bias value 2630. In FIG. 26, the mid-voltagebias value 2630 is 2{circumflex over ( )}11−1, and the DAC 310 is a 12bit DAC.

The first logic chain 2612 provides the positive and negative outputs ofthe first logic chain 2612 to combiners 2624, 2626. The positive andnegative outputs of the first logic chain 2612 correspond to the baseexcitation signal 452 (or a first harmonic of the dithered excitationsignal 352).

The second logic chain 2614 generates the dither to be added to(combined with) the excitation signal generated by the first logic chain2612. The second logic chain 2614 receives two inputs: a high order evenharmonic of the sine function at input 2662 and an amplitude setting forthe high order even harmonic of the sine function at input 2664. In someimplementations, the second logic chain 2614 reduces a number of bits ofthe high order even harmonic of the sine function, the amplitude settingof the high order even harmonic, or both. Additionally or alternatively,the second logic chain 2614 shifts the amplitude setting by addition ofa user defined value, e.g., 0.01 in FIG. 26. The second logic chain 2614multiplies the high order even harmonic of the sine function by theshifted amplitude setting to generate a high order even harmonic signal2666 (i.e., an example of the dither of the dithered excitation signal352).

The second logic chain 2614 generates differential outputs (e.g., apositive output and a negative output) based on the high order evenharmonic signal 2666. For example, the second logic chain 2614 includesconversion logic 2622 configured to generate the differential outputsbased on generating a positive value and a negative value for each valueof the high order even harmonic signal 2666.

The second logic chain 2612 (or conversion logic 2622) may convert thepositive and negative outputs to unsigned values and may further reducea number of bits of the positive and negative outputs for processing bythe DAC 310 similar to the first logic chain 2602.

Operation of the third logic chain 2606 is similar to operation of thesecond logic chain 2614. In the particular implementation, an amplitudesetting of the third logic chain 2602 is zero, i.e., the logic 2602 doesnot use the third harmonic to generate the excitation signal.

The combiners 2624, 2626 combine the outputs of the logic chains2612-2616. For example, the combiner 2624 adds the positive outputs ofthe first logic chain 2612 and the second logic chain 2614 to generate apositive combined output. The combiner 2624 adds the negative outputs ofthe first logic chain 2612 and the second logic chain 2614 to generatenegative combined output. The logic 2602 provides the positive andnegative combined outputs to the DAC 310 via output terminals 2642,2644. The DAC 310 converts the combined outputs into the ditheredexcitation signal 352. In other implementations, the combiners 2624,2626 may subtract the outputs of the third logic chain 2616.

In some implementations, the logic 2602 includes logic for aninitialization process. As illustrated in FIG. 26, the logic 2602includes switches 2632, 2634, a Boolean input 2636 (labeled freeze inFIG. 26), and a mid-voltage initialization value 2638. In the exampleillustrated in FIG. 26, the DAC 310 is a 12 bit DAC and the mid-voltagevalue is half of the 12 bit value (2{circumflex over ( )}11−1). Theswitches 2632, 2634 are configured to switch between outputting thecombined excitation signal output of the combiners 2624, 2626 or themid-voltage initialization value 2638 based on the Boolean input 2636.As illustrated in FIG. 26, the logic 2602 is not operating in theinitialization mode, as indicated by a position of the switches 2632,2634.

In some implementations, the sine function inputs are generated using atable (e.g., a look-up table) of stored values. In a particularimplementation, the look-up table for the sine function inputs isshared, i.e., one sine look-up table that generates the sine functioninputs for the three logic chains 2612-2616. Additionally, the sinelook-up table may be shared with other components and logic of theinertial measurement unit 102, as described further herein. For example,the sine look-up table may be shared with other components and logic viaRound-Robin scheduling. In other implementations, the sine functioninputs are calculated using a series of arithmetic logic operations(e.g., add and shift operations) or Taylor expansion.

FIG. 27 is a circuit diagram 2700 that illustrates an example of aresolver driver circuit 2702. The resolver driver circuit 2702 generatesa differential excitation signal, such as the excitation signals 352,452 of FIGS. 3 and 4) and sends the differential excitation signal to aresolver via terminals 2712 and 2714. The resolver driver circuit 2702may include or correspond to the excitation signal generation system 202of FIG. 2. In some implementations, the differential excitation signalis time coordinated with the current drive switching of the motor.

The resolver driver circuit 2702 includes a low precision ADC 2704 and ahigh precision ADC 2706. As an example, the low precision ADC 2704 maybe a 10 or 12 bit ADC and the high precision ADC 2706 is a 16 bit ADC.In other implementations, other size ADCs may be used. The low precisionADC 2704 and the high precision ADC 2706 are configured to timecoordinate the excitation signal (or the dithered excitation signal 352)with the PWM 242, the ADCs 318, 320, and the demodulation system 222 ofFIGS. 2 and 3.

The low precision ADC 2704 is configured to receive resolver outputs andgenerate the differential excitation signal. The low precision ADC 2704provides the differential excitation signal to the resolver(s). Forexample, the low precision ADC 2704 provides the differential signal toa particular resolver of the dual speed resolver 312 of FIG. 3 via theterminals 2712 and 2714. The low precision ADC 2704 is also configuredto generate serial peripheral interface (SPI) excitation signals 2742and to send the SPI excitation signals 2742 to components of theinertial measurement unit 102, such as an FPGA thereof.

The high precision ADC 2706 is configured to receive resolver outputsand generate SPI feedback signals 2744. The SPI feedback signals 2744are provided to components of the inertial measurement unit 102, such asan FPGA thereof. The SPI excitation and feedback signals 2742, 2744enable time coordination between components of the inertial measurementunit 102. The SPI excitation and feedback signals 2742, 2744 may begenerated based on the resolver feedback. For example, the SPIexcitation signals 2742 are generated based on positive sine and cosinefeedback (and independent of negative sine and cosine feedback) and theSPI feedback signals 2744 are based on the differential sine and cosinefeedback.

During operation, the resolver driver circuit 2702 receives differentialsine feedback from a particular resolver (such as one of the resolvers342, 344) of the dual speed resolver 312 at terminals 2722 and 2724. Theresolver driver circuit 2702 receives differential cosine feedback fromthe particular resolver at terminals 2732 and 2734. The differentialsine and cosine feedback are used to generate the differential resolverexcitation signal and the feedback signal. Because the differentialexcitation signal and the SPI excitation signals 2742 are generatedbased on the resolver feedback, the DAC 310, the dual speed resolver312, the differential voltage sensors 314, 316, or a combinationthereof, can be time coordinated. Because the SPI feedback signals 2744are generated based on the resolver feedback and indicate a timing ofthe resolver system 204 (e.g., a timing of inputs and outputs), thedemodulation system 222, the dual resolver combination system 224, thefeedback control system 232, the PWM 242, or a combination thereof, canbe time coordinated with each other and the components time coordinatedby the SPI excitation signals 2742. To illustrate, the SPI excitationand feedback signals 2742, 2744 indicate data and clock data (e.g., whento sample the data), thereby enabling synchronization betweencomponents. Additionally or alternatively, the PWM 242 can be timecoordinated with the above components by counter or clocksynchronization, as described with reference to FIGS. 33-35.

FIG. 28 is a circuit diagram 2800 that illustrates an example of a motordriver circuit 2802. The motor driver circuit 2802 is coupled to a powersupply and a motor, such as the power supply 252 and the motors 124 ofFIGS. 1 and 2. The motor driver circuit 2802 illustrated in FIG. 28,corresponds to an inverter for a 3-phase motor, such as the inverter 112of FIG. 1. In such implementations, the power supply 252 provides themotor driver circuit 2802 with DC power. The motor driver circuit 2802converts the DC power into an AC signal and provide the AC signal to themotor, such as provides the AC power signal 380 to the motor 124 asdescribed with reference to FIG. 3. In other implementations, the motors124 may have more than three phases or less than three phases. For aninertial measurement unit 102 that includes the gimbal device 114 withmultiple gimbals 122, the inverter 112 may have a motor driver circuit2802 for each motor 124 that drives a corresponding gimbal 122.

The motor driver circuit 2802 includes six transistors 2822-2826,2832-2836 arranged in three half bridges. In other implementations, thesix transistors 2822-2826, 2832-2836 may be arranged in full bridges.The six transistors 2822-2826, 2832-2836 are configured to control powerdelivery to a particular motor 124. First, second, and third transistors2822-2826 correspond to upper transistors, and fourth, fifth, and sixthtransistors 2832-2836 correspond to lower transistors. The first andfourth transistors 2822, 2832 correspond to a first phase (first lane)of the motor. The second and fifth transistors 2824, 2834 correspond toa second phase (second lane) of the motor, and the third and sixthtransistors 2826, 2836 correspond to a third phase (third lane) of themotor. In some implementations, the six transistors 2822-2826, 2832-2836include or correspond to N-channel MOSFETs. The six transistors2822-2826, 2832-2836 may include or corresponds the transistors 336 ofFIG. 3.

In some implementations, the motor driver circuit 2802 includes one ormore high bandwidth current sensors 2812 configured to determine acurrent of the motor driver circuit 2802. As illustrated in FIG. 28, themotor driver circuit 2802 includes two high bandwidth current sensors2812. In other implementations, the motor driver circuit 2802 includesmore than two high bandwidth current sensors 2812 or fewer than two highbandwidth current sensors 2812. In the particular example illustrated inFIG. 28, the high bandwidth current sensors 2812 include or correspondto Hall Effect current sensors. The high bandwidth current sensors 2812are configured to perform continuous built in testing of current of themotor driver circuit 2802.

In some implementations, the motor driver circuit 2802 includes one ormore voltage sensors 2814 configured to determine a voltage of the motordriver circuit 2802. As illustrated in FIG. 28, the motor driver circuit2802 includes three voltage sensors 2814, each voltage sensor coupled toan output terminal (e.g., source) of the upper transistors 2822-2826. Asillustrated in FIG. 28, the voltage sensors 2814 are configured todetermine a voltage of an output of the upper transistors 2822-2826,which may correspond to a voltage provided to the motor. FIG. 28 alsoillustrates a schematic diagram of the first transistor 2822 of theupper transistors 2822-2826. The schematic diagram illustrates threeterminals 2862-2866. The transistor 2822 includes a first terminal 2862(e.g., a drain), a second terminal 2864 (e.g., a gate), and a thirdterminal 2866 (e.g., a source).

The motor driver circuit 2802 includes gate drivers 2842 to drive gates(the second terminals 2864) of the six transistors. The gate drivers2842 are configured to generate and provide activation signals to thegates (the second terminals 2864) of the six transistors 2822-2826,2832-2836. For example, the gate drivers 2842 generate high and lowlogical signals responsive to PWM pulse signals and provide the high andlow logical signal to the gates (the second terminals 2864) of the sixtransistors 2822-2826, 2832-2836, which control the 3-phase powerdelivery to the motor.

The motor driver circuit 2802 may include a Bootstrap power supplycircuitry 2844 for the upper transistors 2822-2826. For example, whenthe upper transistors 2822-2826 include N-channel MOSFETs, the Bootstrappower supply circuitry 2844 is used to drive the upper transistors2822-2826.

The motor driver circuit 2802 is coupled to the motors 124 of FIG. 1 viaoutputs between the upper transistors 2822-2826 and the lowertransistors 2832-2836, such as via output 2852. As illustrated in FIG.28, the third terminals 2866 (e.g., the source) of the upper transistors2822-2826 and the first terminals 2862 (e.g., the drain) of the lowertransistors 2832-2836 are coupled to the motor.

In some implementations, the motor driver circuit 2802 is controlled byan FPGA and is time coordinated with resolver excitation. For example,the motor driver circuit 2802 is controlled by the PWM 242 of FIG. 2, asdescribed with reference to FIGS. 32-35. Additionally or alternatively,the motor driver circuit 2802 is time coordinated with resolver sensing.Time coordination with resolver excitation and sensing reduces noise anderrors in resolver outputs and increases precision of determination of adrive shaft of the motor. For example, time coordination reduceselectromagnetic interference caused by motor drive switching.

FIG. 29 is a diagram 2900 that illustrates an example of cascadedfeedback logic 2902 for speed feedback and position feedback. Thecascaded feedback logic 2902 includes an outer loop 2912 and an innerloop 2914. The outer and inner loops 2912, 2914 refer to feedback loopscorresponding to position and speed feedback 2924, 2936 generated as aresult of a current command 2940. The current command may include orcorresponds to the current command 372 of FIG. 3.

The outer loop 2912 corresponds to a position feedback loop and isconfigured to generate a rate command 2930 based on a received positioncommand 2922. The position command 2922 and the position feedback 2924are used to generate an intermediary signal (e.g., an error signal).Control gain is applied to the intermediary signal to generate the ratecommand 2930. The rate command 2930 indicates a rate, such as a rate ofchange in position. The rate command 2930 (generated by the cascadedfeedback logic 2902) may indicate a speed similar to a speed command2932 (received from a flight computer), but the rate command 2930 may bein different units than the speed command 2932. For example, the ratecommand 2930 may be in radians per second and the speed command 2932 maybe in RPM.

The inner loop 2914 is configured to limit the rate command 2930 basedon the speed command 2932 (e.g., an RPM command) to generate a limitedrate command 2934. The speed feedback 2936 (e.g., RPM feedback) isapplied to the limited rate command 2934 and second control gain, suchas an RPM gain 2938 (Krpm), is applied to generate the current command2940. The inner loop 2914 outputs the current command 2940 and thecurrent command 2940 is converted into a duty cycle value for the PWM242, such as by the current tracker 330 of FIG. 3.

During operation, the outer loop 2912 receives the position command 2922from the flight computer 254 of FIG. 2 and the position feedback 2924from the resolver system 204. The outer loop 2912 generates an errorsignal based on subtracting the position feedback 2924 from the positioncommand 2922. The control gain is applied to the error signal togenerate the rate command 2930. For example, the error signal ismultiplied by a proportional gain 2926 and the error signal (or anintegral thereof) is multiplied by an integral gain 2928. The outer loop2912 provides the rate command 2930 is to the inner loop 2914.

The inner loop 2914 rate limits the rate command 2930 based on the speedcommand 2932. For example, the rate command 2930 is decreased when thespeed command 2932 indicates a lower RPM than the rate command 2930. Toillustrate, when the RPM indicated by the speed command 2932 correspondsto a lesser (slower) change in position value than the change inposition value of the rate command 2930, the inner loop 2914 decreasesthe change in position value of the rate command 2930 to the change inposition value of the speed command 2932. The inner loop 2914 subtractsthe speed feedback 2936, received from the resolver system 204 of FIG.2, from the limited rate command 2934 to generate a second error signal.The current command 2940 is generated based on applying second controlgain to the second error signal. For example, the second error signal ismultiplied by the RPM gain 2938. The inner loop 2914 provides thecurrent command 2940, indicative of torque of the motor 124 of FIG. 1,to the inverter 112 of FIG. 1, which controls operation of the motor 124based on the current command 2940. The inner loop 2914 of the cascadedfeedback logic 2902 can be difficult to implement, especially for lowdamped systems, such as the motors 124 used to position the gimbals 122of FIG. 1.

FIG. 30 is a logic diagram 3000 that illustrates an example of logic3002 for combined speed and position feedback control. As compared tothe diagram 2900 of FIG. 29, the logic 3002 includes a single feedbackloop for combined position and speed feedback. As opposed to the diagram2900, the logic 3002 uses the speed feedback 2936 for damping, not as aseparate feedback loop. Additionally, the logic 3002 generates a ratelimited position command 3034 (e.g., a combined speed and positioncommand) before applying feedback. The logic 3002 provides improvedcontrol and lower complexity for low damped motors, such as the motors124 used to position the gimbals 122 of FIG. 1, as compared to thecascaded feedback logic 2902 of FIG. 29.

During operation, the logic 3002 receives the position command 2922 andthe speed command 2932 from the flight computer 254 of FIG. 2. The logic3002 rate limits the position command 2922 based on the speed command2932 to generate the rate limited position command 3034. For example, aposition value of the position command 2922 is decreased when the speedcommand 2932 (converted into a position value representing a maximumchange in position relative to a previous position) indicates a lesserchange in position relative to the previous position than a secondchange of position relative to the previous position indicated by theposition value of the position command 2922.

The rate limited position command 3034 is generated before an errorsignal 3012 is generated (or control gain is applied). The logic 3002generates the error signal 3012 by subtracting the position feedback2924 from the rate limited position command 3034. For example, theposition value of the rate limited position command 3034 is adjustedbased on the position value of the position feedback 2924 to generate aposition error.

The logic 3002 applies control gain to the error signal 3012. Forexample, the logic 3002 multiplies the error signal 3012 by theproportional gain 2926 and multiplies the error signal 3012 by theintegral gain 2928. The logic 3002 generates an adjusted error signal3014 based on a sum of the two products of the error signal 3012 andgains 2926, 2928. In a particular implementation, the first productcorresponds to a derivative of the error signal 3012 multiplied by theproportional gain 2926.

Additionally, the logic 3002 may dampen the adjusted error signal 3014based on the speed feedback 2936. The logic 3002 multiplies the speedfeedback 2936 by the RPM gain 2938 (e.g., a damping factor) to generatean RPM damping value 3016. The logic 3002 subtracts the RPM dampingvalue 3016 from the adjusted error signal 3014 to dampen the adjustederror signal 3014. For example, damping the adjusted error signal 3014generates a damped error signal. The logic 3002 generates the currentcommand 2940 based on the damped error signal. For example, the logic3002 integrates the damped error signal to generate the current command2940. As compared to the cascaded feedback logic 2902 of FIG. 29, thelogic 3002 provides improved control and stability for low damped motorswith lower complexity (e.g., without multiple or nested control loops).

FIGS. 31 and 32 are logic diagrams that illustrate examples of logic forcombined speed and position feedback including multiple operating modes.As illustrated in FIGS. 31 and 32, the diagrams have been simplified forclarity. For example, some logic boxes of the diagram of FIG. 31 areblank and are discussed with respect to FIG. 32 and vice versa.Additionally, some of the conversions, shifts, holds and constant valueshave been omitted from the description for clarity. In FIGS. 31 and 32,conversions logic boxes are represented by the letter “C” for bitconversions and “si” for signed/unsigned conversions, shift logic boxesare represented by a three character string including the letter “S”, adirection (i.e., “L” or “R”), and a number of bits, absolute value logicboxes are represented by “abs”, and holds or delay units are representedby “1/z”.

Referring to FIG. 31, a logic diagram 3100 of an example of logic 3102for combined speed and position feedback control including a directspeed command mode is illustrated. As compared to the logic 3002 of FIG.30, the logic 3102 is configured to operate in a direct speed mode or ina position and speed mode. By utilizing both the direct speed mode andthe position and speed mode, the logic 3102 can control the motor 124 byadjusting torque or speed of the motor 124. The logic 3102 includesdirect speed command mode logic 3112, position command logic 3114, speedcommand logic 3116, rate limiter logic 3118, error generation logic3120, combined proportional gain logic 3122, combined integral gainlogic 3124, and output logic 3126.

The direct speed command mode logic 3112 is configured to generate aderived position command 3132 based on the speed command 2932. Thederived position command 3132 indicates an incremental change inposition from a previous position. The direct speed mode may include orcorrespond to a direct RPM command mode. The direct RPM command modeincludes or corresponds to a mode where the flight computer 254 providesthe speed command 2932 (RPM command) only in the commands 370 of FIG. 3.After switching from a first mode (a position and speed mode) to asecond mode (a direct speed mode), the direct speed command mode logic3112 is configured to output a position (e.g., the estimated position366) of the motor determined by the resolver system 204 of FIG. 2instead of the derived position command 3132. By outputting theestimated position 366 after switching modes, the direct speed commandmode logic 3112 enables smooth operation of the motor (e.g., reduces oreliminates jerks from switching modes). Operations of the direct speedcommand mode logic 3112 may be performed by direct speed circuitry. Asan example, the direct speed circuitry includes one more combiners,multipliers, delay elements, switches, etc., to perform the operationsof the direct speed command mode logic 3112, as illustrated in FIG. 31.

The position command logic 3114 is configured to receive the positioncommand 2922, such as from the flight computer 254 of FIG. 2, and tooutput the position command 2922 to the rate limiter logic 3118 when inthe first mode. The position command logic 3114 is configured to outputthe derived position command 3132 when in the second mode. For example,the position command logic 3114 includes a switch 3134 configured toprovide the derived position command 3132 to the rate limiter logic 3118based on a Boolean input indicating that the logic 3102 is operating inthe second mode. Accordingly, the output logic 3126 generates thecurrent command 2940 based on the speed command 2932 and independent ofa position command 2922 when operating in the second mode.

The speed command logic 3116 is configured to receive the speed command2932 and to provide the speed command 2932 to the rate limiter logic3118. In some implementations, the speed command logic 3116 isconfigured to convert the speed command 2932 into an unsigned integerand to convert a value of the speed command 2932 into an absolute value.Additionally or alternatively, the speed command 2932 may be adjustedfor a resolver. For example, the speed command 2932 may be adjusted bymultiplying the speed command 2932 by a constant value to account for adifference between the RPM of the motor and resolver speed.

The rate limiter logic 3118 is configured to generate the rate limitedposition command 3034 based on the position command 2922 and the speedcommand 2932 when in the first mode. The rate limiter logic 3118 isconfigured to generate the rate limited position command 3034 based onthe speed command 2932 when in the second mode. In some implementations,the rate limiter logic 3118 is further configured to operate in aninitialization mode, as described with reference to FIG. 32.

The error generation logic 3120 is configured to generate the errorsignal 3012 (indicative of a position error) based on the rate limitedposition command 3034 and the position feedback 2924 of the resolversystem 204 of FIG. 2. For example, the error generation logic 3120generates the error signal 3012 by subtracting the position feedback2924 from the rate limited position command 3034. Operations of theerror generation logic 3120 may be performed by error signal generationcircuitry. As an example, the error signal generation circuitry includesa combiner, as illustrated in FIG. 31.

The combined proportional gain logic 3122 is configured to applyproportional gain 2926 to the error signal 3012. The combinedproportional gain logic 3122 is configured to output a product of theerror signal 3012 and the proportional gain 2926 to the output logic3126. In a particular implementation, the proportional gain 2926 ismultiplied by a derivative of the error signal 3012 to generate theproduct. The combined integral gain logic 3124 is configured to applyintegral gain 2928 to the error signal 3012. The combined integral gainlogic 3124 is configured to output a product of the error signal 3012and the integral gain 2928 to the output logic 3126.

The output logic 3126 is configured to generate the current command 2940based on the proportional gain 2926 and the integral gain 2928. Forexample, the output logic 3126 is configured to add the two products togenerate the adjusted error signal 3014. The output logic 3126 isconfigured to generate the current command 2940 based on the adjustederror signal 3014. For example, the output logic 3126 is configured tointegrate the adjusted error signal 3014 to generate the current command2940. In a particular implementation, the integrated adjusted errorsignal 3014 may be limited based on a maximum current value and aminimum current value to generate the current command 2940. The outputlogic 3126 outputs the current command 2940 via an output terminal 3130.

In some implementations, the logic 3102 further includes damping logic3128. The damping logic 3128 generates the damping value 3016 which isapplied to the adjusted error signal 3014 to generate a damped errorsignal 3138. In such implementations, the damped error signal 3138 isintegrated to generate the current command 2940. In addition, the logic3102 may include a switch (not shown) to control a damping mode, i.e.,when damping is applied or not. Operations of the damping logic 312 maybe performed by damping circuitry. As an example, the damping circuitryincludes one more combiners, multipliers, delay elements, switches,etc., to perform the operations of the damping logic 3128, asillustrated in FIG. 31.

Referring to FIG. 32, a logic diagram 3200 of an example of logic 3202for combined speed and position feedback control including aninitialization mode is illustrated. The logic 3202 includes similarlogic to the logic 3102 of FIG. 31 and additionally includesinitialization logic 3204. The initialization logic 3204 includestracking logic 3212 and initial condition (ic) logic 3214.

The initialization logic 3204 is configured to receive and generateinitialization inputs. For example, the tracking logic 3212 isconfigured to receive a position tracking enable input 3222 and an RPMtracking enable input 3224. The tracking logic 3212 generates a trackinginput 3226 as an output based on either the position tracking enableinput 3222 or the RPM tracking enable input 3224 indicating true (ortracking enabled).

The ic logic 3214 is configured to generate an ic input 3232 as anoutput based on the position tracking enable input 3222, the RPMtracking enable input 3224, an initialization filter input 3228, theservo offset output 1866, and an RPM pulse tracking enable input 3230.

The logic 3202, as compared to the logic 3102, includes additionalswitches 3242-3248. The switches 3422-3428 are configured to inputinitialization values (e.g., 0 or null values) responsive to one or moreof the initialization inputs 3222-3232. For example, a first switch 3242receives the initialization filter input 3228 and outputs a position ofthe motor determined by the resolver or outputs the position command2922 based on the initialization filter input 3228. To illustrate, whenin initialization mode, the estimated position 366 of the motor isprovided to the rate limiter logic 3118, and after initialization mode,the position command 2922 is provided to the rate limiter logic 3118.

Additionally, the rate limiter logic 3118 is further configured toreceive a present or current position (e.g., the estimated position 366of FIG. 3) of the motor determined by the resolver system 204 of FIG. 2and an ic enable input. The rate limiter logic 3118 is configured tooutput the present position of the motor as the rate limited positioncommand 3034 responsive to the ic enable input indicating true. In aparticular implementation, the ic enable input is generated based on theic input 3232 and a shutdown input 3236. For example, an OR logic gategenerates the ic enable input based on the ic input 3232 and theshutdown input 3236. To illustrate, when either of the ic input 3232 orthe shutdown input 3236 indicates true, the ic enable input indicatestrue and the rate limiter logic 3118 outputs the present position (e.g.,the estimated position 366) of the motor determined by the resolver asthe rate limited position command 3034.

A second switch 3244 receives the tracking input 3226 and outputs a zerovalue or the adjusted error signal 3014 (or the damped error signal3138) based on the tracking input 3226. A third switch 3246 receives afreeze input 3234 and outputs a zero value or the adjusted error signal3014 (or the damped error signal 3138) based on the freeze input 3234.Thus, if either the tracking input 3226 or the freeze input 3234 aretrue, the third switch 3246 outputs a zero instead of the adjusted errorsignal 3014 (or the damped error signal 3138).

A fourth switch 3248 receives a Boolean output generated based on the icinput 3232 and the shutdown input 3236. Based the Boolean output, thefourth switch 3248 outputs a zero value or the current command. In theexample illustrated in FIG. 32, the Boolean output is output by an ORlogic gate, thus if either the ic input 3232 or the shutdown input 3236are true, the switch 3248 outputs a zero. To illustrate, the logic 3202outputs the current command 2940 indicating zero current duringinitialization and shutdown modes.

In other implementations, equivalent logic may be used. For example, thesecond and third switches 3244 and 3246 may be replaced by a singleswitch and a logic gate, similar to the fourth switch 3248 andcorresponding OR logic gate.

FIG. 33 is a diagram 3300 that illustrates an example of PWM operationwith an adjustable comparison criterion. A PWM, such as the PWM 242 ofFIG. 2, receives a set point signal that includes a comparison value andat least one comparison criterion. The at least one comparison criterionincludes a Boolean condition. For example, the Boolean conditionincludes or corresponds to a greater than condition, a less thancondition, a greater than or equal to condition or a less than or equalto condition. In other implementations, the comparison value and the atleast one comparison criterion are received in separate signals. The PWM242 is configured to control power delivery to a controlled component.The controlled component may include a light emitting diode, a dutycycle controller, a clock signal generator, a buck converter, a servo, astepper motor, a single phase motor, or a multi-phase motor.

In FIG. 33, a counter 3302, such as an up-down counter, is configured tocount up from zero to a particular number and then back down from theparticular number to zero. As illustrated in FIG. 33 the counter 3302counts up to 8192 and a counter signal 3304 generated by the counter3302 is depicted by the dashed line. The PWM 242 includes a comparatorto compare a counter value of the counter signal 3304 to a comparisonvalue 3322 (CMP) indicated by a set point signal. In someimplementations, the counter 3302 is synchronized with the counter 842of FIG. 8. Synchronization of the counters 842 and 3302 enables currentdrive switching of the motor to be synchronized with (e.g., offset from)excitation signals of the resolver system 204 of FIG. 2 and enablesmasking data affected by current drive switching.

FIG. 33 depicts operation of the PWM 242 for five different set pointsignals 3312-3320. A first set point signal 3312 includes a comparisonvalue 3322 and two comparison criteria 3324, 3326. The PWM 242determines the comparison value 3322 and the two comparison criteria3324, 3326 by processing the set point signals 3312-3320. For example,the PWM 242 right shifts the first set point signal 3312 to get thecomparison value 3322. The PWM 242 uses the two least significant bits(LSBs) of the first set point signal 3312 to determine the twocomparison criteria 3324, 3326. As illustrated in FIG. 1, the first setpoint signal 3312 has a value of 16384. Right shifting the value 16384of the first set point signal 3314 generates a value of 4096 for thecomparison value 3322. In binary, the two LSBs of 16384 are “00.” ThePWM 242 determines that a two bit value of “00” corresponds to a firstcomparison criterion 3324 of greater than and a second comparisoncriterion 3326 of greater than. The PWM 242 may determine that the twobit value corresponds to or indicates the two comparison criteria 3324,3326 by processing the two bit value with logic or by performing a tablelookup.

The PWM 242 provides a high signal (or causes a high signal to beprovided) to a gate of a transistor responsive to the comparison value3322 satisfying the comparison criteria 3324, 3326 with respect to avalue of the counter 3302. Providing the high signal to the transistoractivates the transistor. As illustrated in FIG. 33, the PWM 242activates the transistor when the comparison value 3322 of 4096 isgreater than value of the counter 3302 (e.g., when the counter valuedecreases below the comparison value 3322 of 4096). In FIG. 33, thetransistor is activated at a counter value of 4095 for the first setpoint signal 3312. Activating the transistor generates a pulse edge of apulse of a PWM signal based on the first set point signal 3312.

The PWM 242 provides a low signal (or causes a low signal to beprovided) to the gate of the transistor responsive to the comparisonvalue 3322 not satisfying the comparison criteria 3324, 3326 withrespect to a value of the counter 3302. Providing the low signal to thetransistor deactivates the transistor. The PWM 242 deactivates thetransistor when comparison value 3322 of 4096 is less than value of thecounter 3302 (e.g., when the counter value increases above thecomparison value 3322 of 4096). In FIG. 33, the transistor isdeactivated at a counter value of 4096 for the first set point signal3312, which means the transistor is on when the counter value is 4095.The on time of the transistor (and the pulse-width of the pulse of thePWM signal) is shown by the area of the triangle formed in FIG. 33 withcross hatching and corresponds to a period of time when the comparisonvalue 3322 is greater than a value of the counter 3302 for the first setpoint signal 3312. The transistor on time of the first set point signal3312 is 8190 counts, i.e., counts corresponding to counter values of4095 to 0 and back from 0 to 4095.

The PWM 242 receives a second set point signal 3314. The second setpoint signal 3314 has a value of 16385. Right shifting the value 16385of the second set point signal 3314 generates a value of 4096 for thecomparison value 3322. In binary, the two LSBs of 16385 are “01.” ThePWM 242 determines that a two bit value of “01” corresponds to a firstcomparison criterion 3324 of greater than or equal to and a secondcomparison criterion 3326 of greater than. As illustrated in FIG. 33,the PWM 242 activates the transistor when the comparison value 3322 of4096 is equal to the value of the counter 3302, i.e., at a counter valueof 4096. Activating the transistor generates a pulse edge of a secondpulse of the PWM signal based on the second set point signal 3314. ThePWM 242 deactivates the transistor when the comparison value 3322 of4096 is equal to (no longer greater than) the value of the counter 3302.In FIG. 33, the PWM 242 deactivates the transistor at a counter value of4096. Deactivating the transistor generates a second pulse edge of thesecond pulse of the PWM signal based on the second set point signal3314. The on time of the transistor (and the pulse-width of the pulse ofthe PWM signal) is shown by the area of the triangle formed in FIG. 33with cross hatching. As compared to the on time of the first set pointsignal 3312, the on time of the second set point signal 3314 is a singlecount (i.e., 1 of 8192) higher. To illustrate, the second set pointsignal 3314 indicates a transistor on time of 8191 counts i.e., countscorresponding to counter values of 4096 to 0 and 0 to 4095.

In conventional PWMs, the comparison criterion is constant (e.g., onlygreater than or only greater than or equal to). Thus, to increase thepulse-width of a pulse the comparison value is changed (similar to thecomparison value changing from 4096 to 4097 between the first and fifthset point signals 3312, 3320). Changing the comparison value 3322increases the on time or pulse-width by two counts. For example,increasing the comparison value 3322 from 4096 to 4097 (with a constantcomparison criterion of greater than) increases the on time count by twofrom 8190 counts for 4096 to 8192 counts for 4097. To illustrate, forthe comparison value 3322 of 4097 the transistor on time is on from 4097to 0 and back from 0 to 4097. Accordingly, the PWM 242 has reducedgranularity and finer precision and control of the motor 124.

The PWM 242 receives a third set point signal 3316. The third set pointsignal 3316 has a value of 16386. Right shifting the value 16385 of thesecond set point signal 3314 generates a value of 4096 for thecomparison value 3322. In binary, the two LSBs of 16386 are “10.” ThePWM 242 determines that a two bit value of “10” corresponds to a firstcomparison criterion 3324 of greater than or equal to and a secondcomparison criterion 3326 of greater than. The third set point signal3316 operates the same as the second set point signal 3314 to generate athird pulse of the PWM signal based on the third set point signal 3316.In multi lane implementations, such as described with reference to FIG.34, the third set point signal 3316 may increase power delivery to acontrolled component in combination with adjustable comparison criterionof a second lane by 1 count.

The PWM 242 receives a fourth set point signal 3318. The fourth setpoint signal 3318 has a value of 16387. In binary, the two LSBs of 16387are “11.” The PWM 242 determines that a two bit value of “11”corresponds to a first comparison criterion of 3324 greater than orequal to and a second comparison criterion 3326 of greater than or equalto. As illustrated in FIG. 33, the PWM 242 activates the transistor whenthe comparison value 3322 of 4096 is equal to the value of the counter3302, i.e., at a counter value of 4096. Activating the transistorgenerates a pulse edge of a fourth pulse of the PWM signal based on thefourth set point signal 3318. The PWM 242 deactivates the transistorwhen the comparison value 3322 of 4096 is less than (no longer equal toor greater than) the value of the counter 3302, i.e., at a counter valueof 4097, which means the transistor is on when the counter value is4096. Deactivating the transistor generates a second pulse edge of thefourth pulse of the PWM signal based on the fourth set point signal3318. As compared to the on time of the second set point signal 3314,the on time of the fourth set point signal 3318 is a single count (i.e.,1 of 8192) higher. To illustrate, the second set point signal 3314indicates a transistor on time of 8191 counts and the fourth set pointsignal 3318 indicates a transistor on time of 8192 counts i.e., countscorresponding to counter values of 4096 to 0 and 0 to 4096.

The PWM 242 receives a fifth set point signal 3320. The fifth set pointsignal 3320 has a value of 16388. In binary, right shifting 16388 by 2gives the comparison value 3322 a value of 4097. The two LSBs of 16388are “00.” The PWM 242 determines that a two bit value of “00”corresponds to a first comparison criterion 3324 of greater than and asecond comparison criterion 3326 of greater than. The fifth set pointsignal 3320 operates with a different comparison value 3322 anddifferent comparison criteria 3324, 3326 as compared to the fourth setpoint signal 3318 but produces the same number of counts of on time asthe fourth set point signal 3318. In other implementations, each bit ofthe two LSBs may corresponds to the particular comparison rule orcondition. For example, a bit having a value of 0 may indicate a greaterthan condition and a bit having a value of 1 may indicate greater thanor equal to condition.

As illustrated in FIG. 33, the on time increases by 0.01250 microsecondsfor adjusting a comparison criterion, as opposed to 0.0250 microsecondsfor adjusting a comparison value. Accordingly, the PWM 242 has reducedgranularity and increased precision and control by adjusting thecomparison criteria 3324, 3326. In the example illustrated in FIG. 33,the PWM 242 has a frequency of 40 Megahertz. A frequency of the resolversystem 204 (2441.2 Hertz) is equal to the frequency of the PWM 242divided by double a period (8192) of the PWM 242.

In other implementations, a single adjustable comparison criterion maybe used and may be indicated by 1 LSB of a set point signal. In suchimplementations, the PWM 242 still has a granularity of 1 counter valueor 0.01250 microseconds per each 1 value adjustment of the set pointsignal. As an example illustration, in response to an increase in avalue of the first set point signal 3312 by one, the PWM 242 wouldoperate similar to the second set point signal 3314. In response to anincrease in a value of the first set point signal 3312 by two, the PWM242 would operate similar to the fifth set point signal 3320 (as opposedto the third set point signal 3316). Accordingly, as compared to rightshifting the set point signal by two for two adjustable comparisoncriteria, the set point signal with a single adjustable comparisoncriterion is right shifted by one. Additionally, the set point signalwith one adjustable comparison criterion would have a range of valuesfrom 0 to 16384, as opposed to a range of values from 0 to 32768 for aset point signal with two adjustable comparison criteria. Suchimplementations may be implemented where multilane control is not used,such as LEDs, one phase motors, etc.

As illustrated in FIG. 33, the set point signals 3312-3320 indicate achange each cycle to illustrated operation of the PWM 242. Duringoperation of a controlled component, the PWM 242 may receive a set pointsignal each cycle, and the set point signal may or may not indicate achange in duty cycle of the controlled component from a previous setpoint signal for a previous cycle. Alternatively, the PWM 242 mayreceive a set point signal indicating a different duty cycle in responseto changes in a duty value and may maintain a current set point signal(i.e., the comparison value 3322 and comparison criteria thereof 3324,3326) until a new set point signal is received.

FIG. 34 is a diagram 3400 that illustrates an example of two lane PWMoperation with an adjustable comparison criterion. Diagram 3400illustrates operation of an A-lane and a B-lane. In a particularimplementation, the A-lane corresponds to an upper transistor of a firstphase and the B-lane corresponds to a lower transistor of a secondphase. For example, the upper transistor corresponds to the 2822 of FIG.28 and the lower transistor corresponds to the 2834 of FIG. 28.

The A-lane operates similar to the PWM operation described with respectto FIG. 33. The B-lane operates similarly to the A-lane, but in reverse.The B-lane is low (e.g., connected to ground) when a particularcomparison value 3422 (CMPB) satisfies the comparison criteria 3424,3426 for the B-lane (e.g., is less than or less than or equal to) withrespect to a value of the counter 3302 of FIG. 33.

When the A-lane is high and the B-lane is low, a voltage difference iscreated across a controlled component, such as a motor, and current isprovided to the controlled component. The A-lane being high and theB-lane being low can be achieved by closing or activating the uppertransistor 2822 and the lower transistor 2834. Activating thetransistors 2822 and 2834 allows current to flow from the battery (orother power source), through the upper transistor 2822 to the motor 124,across the motor 124 to the lower transistor 2834, and through the lowertransistor 2834 to ground. As compared to the pulses of the PWM 242described with respect to FIG. 33, pulses of the PWM 24 for the A and Blanes are generated by a combination of a set point signal of each lane.Said another way, each set point signal of a particular lane generates apulse edge of two different pulses (rather than two pulse edges of asingle pulse as in FIG. 33). In FIG. 33, pulses provided to a controlledcomponent (e.g., a motor) corresponded to cross hatched triangles. InFIG. 34, pulses provided to a controlled component (e.g., a motor)correspond to dashed rectangles. In the implementation illustrated inFIG. 34, each set point signal of the set point signals 3312-3320generates a first pulse edge of a first pulse and a second pulse edge ofa second pulse and a corresponding set point signal of the set pointsignals 3412-3420 generates a second pulse edge of the first pulse and afirst pulse edge of the second pulse.

The B-lane receives a first set point signal 3412. The first set pointsignal 3412 has a value of 16384. Right shifting the first set pointsignal 3412 value of 16384 by two (i.e., dividing by 4) indicates thecomparison value 3422 (CMPB) of 4096 for the B-lane. The two LSB's of16384 are “00,” and for the B-lane the two LSB's indicate a firstcomparison criterion 3424 of less than or equal to and a secondcomparison criterion 3426 of less than or equal to. The B-lane is lowwhile the comparison value 3422 is less than or equal to a value of thecounter 3302, as illustrated in FIG. 34 by horizontal hatching.

The PWM 242 provides a low signal (or causes a low signal to beprovided) to the gate of the lower transistor 2834 responsive to thecomparison value 3422 not satisfying the comparison criteria 3424, 3426with respect to a value of the counter 3302. Providing the low signal tothe lower transistor 2834 deactivates the lower transistor 2834. The PWM242 deactivates the lower transistor 2834 when comparison value 3422 of4096 is less than or equal to the value of the counter 3302 (e.g., whenthe counter value decreases below the comparison value 3322 of 4096). InFIG. 34, the lower transistor 2834 is deactivated at a counter value of4095 for the first set point signal 3412, which means the lowertransistor 2834 is on when the counter value is 4096. Activating theupper transistor 2822 generates a first pulse edge of a first pulse ofthe PWM signal based on the first set point signals 3312, 3412, anddeactivating the lower transistor 2834 generates a second pulse edge ofthe first pulse of the PWM signal based on the first set point signals3312, 3412.

The on time of the lower transistor 2834 is shown by the area of thetriangle formed in FIG. 34 with cross hatching and corresponds to aperiod of time when the comparison value 3422 is less than or equal tothe value of the counter 3302 for the first set point signal 3412. Theon time of the lower transistor 2834 is indicated by the first set pointsignal 3412 and a previous set point signal (not shown). The set pointsignals 3412-3420 correspond to an off time of the lower transistor2834, and the set point signals 3312-3320 correspond to an on time ofthe upper transistor 2822.

The PWM 242 provides a high signal (or causes a high signal to beprovided) to a gate of the lower transistor 2834 responsive to thecomparison value 3422 satisfying the comparison criteria 3424, 3426 withrespect to a value of the counter 3302. Providing the high signal to thelower transistor 2834 activates the lower transistor 2834. Asillustrated in FIG. 34, the PWM 242 activates the lower transistor 2834when the comparison value 3422 of 4096 is less than or equal to thevalue of the counter 3302 (e.g., when the counter value reaches thecomparison value 3422 of 4096). In FIG. 34, the lower transistor 2834 isactivated at a counter value of 4096 for the first set point signal3412. Activating the lower transistor 2834 generates a first pulse edgeof a second pulse of the PWM signal based on the first set point signals3312, 3412, and deactivating the upper transistor 2822 generates asecond pulse edge of the second pulse of the PWM signal based on thefirst set point signals 3312, 3412.

The first and second pulses of the PWM signal indicate a motor on timeand correspond to overlapping of the on time of the A and B lanes (crosshatched triangles). The first and second pulses are indicated byrectangles in dashed lines in FIG. 34. As shown in FIG. 34, a firstmotor on time associated with the first pulse of the first set pointsignals 3312, 3412 is 25 ns and a second motor on time associated withthe second pulse of the first set point signals 3312, 3412 is 25 ns.Thus, a total motor on time associated with the first set point signals3312, 3412 of the A and B lanes is 50 nanoseconds.

The B-lane receives the second set point signal 3414 having a value of16383. Right shifting the second set point signal 3414 value of 16383 bytwo (i.e., dividing by 4) indicates the comparison value 3422 of 4095.The two LSB's of 16383 are “11,” and for the B-lane the two LSB'sindicate a first comparison criterion 3424 of less than and a secondcomparison criterion 3426 of less than. Because the comparison criteria3424, 3426 changed and the comparison value 3422 changed, the time theB-lane was off did not change. However, because one comparison criterion(i.e., the first comparison criterion 3324) of the A-lane changed in thecorresponding second set point signal 3314, the total motor on time(duty cycle) increased by 25 ns for the second set point signals 3314,3414 with respect to the previous total motor on time of 50 ns for thefirst set point signals 3312, 3412. As shown in FIG. 34, a first motoron time associated with the first pulse of the second set point signals3314, 3414 is 50 ns and a second motor on time associated with thesecond pulse of the first set point signals 3314, 3414 is 25 ns, for atotal motor on time of 75 ns.

The B-lane receives the third set point signal 3416 having a value of16382. Right shifting the third set point signal 3416 value of 16382 bytwo (i.e., dividing by 4) indicates the comparison value 3322 of 4095.The two LSB's of 16382 are “10,” and for the B-lane the two LSB'sindicate a first comparison criterion 3424 of less than and a secondcomparison criterion 3426 of less than or equal to. The A-lane did notchange comparison criteria 3324, 3326 or comparison value 3322 from thesecond set point signal 3312, as explained with reference to FIG. 33.Because the second comparison criterion 3426 of the second set pointsignal 3416 changed on the B-lane, the time the B-lane was off increasedand the motor 124 on time (duty cycle) increased by 25 ns with respectto the previous total motor on time of 75 ns for the second set pointsignals 3314, 3414. As shown in FIG. 34, first and second motor on timesof the third set point signals 3316, 3416 are 50 ns, for a total motoron time of 100 ns.

The B-lane receives the fourth set point signal 3418 having a value of16381. Right shifting the fourth set point signal 3418 value of 16381 bytwo (i.e., dividing by 4) indicates a comparison value of 4095. The twoLSB's of 16381 are “01,” and for the B-lane the two LSB's indicate afirst comparison criterion 3424 of less than and a second comparisoncriterion 3426 of less than or equal to. Because the comparison criteria3424, 3426 changed and the comparison value 3422 changed, the time theB-lane was low did not change. However, because one comparison criterion(i.e., the second comparison criterion 3326) of the A-lane changed inthe corresponding fourth set point signal 3318, the motor 124 on time(duty cycle) increased by 25 ns with respect to the previous total motoron time of 100 ns for the third set point signals 3316, 3416. As shownin FIG. 34, a first motor on time associated with the first pulse of thefourth set point signals 3318, 3418 is 75 ns and a second motor on timeassociated with the second pulse of the fourth set point signals 3318,3418 is 50 ns, for a total motor on time of 125 ns.

The B-lane receives the fifth set point signal 3420 having a value of16381. Right shifting the fifth set point signal 3420 value of 16380 bytwo (i.e., dividing by 4) indicates a comparison value 3422 of 4095. Thetwo LSB's of 16380 are “00,” and for the B-lane the two LSB's indicate afirst comparison criterion 3424 of less than or equal to and a secondcomparison criterion 3426 of less than or equal to. Because the firstcomparison criterion 3424 changed on the B-lane, the time the B-lane waslow increased and the motor 124 on time (duty cycle) increased by 25 nswith respect to the previous total motor on time of 125 ns for thefourth set point signals 3318, 3418. As shown in FIG. 34, a first motoron time associated with the first pulse of the fifth set point signals3320, 3420 is 75 ns and a second motor on time associated with thesecond pulse of the fifth set point signals 3320, 3420 is 75 ns, for atotal motor on time of 150 ns.

As illustrated in FIG. 34, the PWM 242 receives multiple set pointsignals, one for each lane. For example, the A-lane receives the secondset point signal 3314 indicating a value 16385 and the B-lane receivesthe second set point signal 3414 indicating a value of 16383. In otherimplementations, the PWM 242 receives a single set point signalindicating a duty cycle value 374 and at least one comparison criterion.The PWM 242 calculates the comparison values 3322, 3422 for each lanebased on the duty cycle value 374. For example, the PWM 242 calculatesthe comparison values 3322, 3422 for each lane using the equationsCMPA=DutyCycle*CTRPRD*4 and CMPB=(1−DutyCycle)*CTRPRD*4, where CTRPRD isthe maximum counter value (8192).

In some implementations, the PWM 242 is configured to control twotransistors of each lane. In a particular implementation, the PWM 242 isconfigured to provide dead band control, i.e., the PWM 242 delaysactivation of a particular transistor of a lane to prevent bothtransistors of the lane from being active at the same time. For example,the PWM 242 delays activation of an upper transistor of a particularlane to allow a lower transistor of the particular lane to deactivate.This delay is referred to as dead time or dead ticks. The PWM 242 isconfigured to activate the transistors further based on a dead tickssetting or a determined number of dead ticks, as described withreference to FIG. 35.

FIG. 35 is a logic diagram 3500 that illustrates an example of logic3502 for a PWM with an adjustable comparison criterion and dead bandcontrol. The PWM logic 3502 is configured to control the transistors 336of FIG. 3 (e.g., transistors 2822-2826 and 2832-2836) and provide powerto the motor 124 of FIG. 1. In FIG. 35, the PWM logic 3502 controls a3-phase motor, with phases or lanes A, B, and C. The PWM logic 3502 isconfigured to determine when to activate the transistors 336 based on acomparison value and at least one comparison criterion, as describedwith reference to FIGS. 33 and 34. In FIG. 35, the PWM logic 3502 isconfigured to determine when to activate the transistors 336 furtherbased on a dead time input 3512 and a drive enable input 3514. Forexample, the PWM logic 3502 may adjust the on timings of the lanes toprovide a dead band such that both the upper and lower transistors of aparticular lane are not on at the same time.

The PWM logic 3502 includes a comparison logic block 3504 for each lane.Each of the comparison logic blocks 3504 is configured to generate twooutputs: a high activation output 3506 and a low activation output 3508,based further on the dead time input 3512 and the drive enable input3514. The high and low activation outputs 3506, 3508 include Booleanoutputs, such as true or false indicated by a logical 0 or 1. The highand low activation outputs 3506, 3508 may include or correspond to deadband activation outputs or adjusted activation outputs as compared tothe on time determined in FIGS. 33 and 34. Providing dead band controlmay reduce a total on time indicated by a particular set point signal,but dead band control does not increase or reduce the granularity of theon time control or a precision of a set point signal indicating anadjustable comparison criterion.

The dead time input 3512 (DT) is configured to adjust the on timings ofthe transistors 336. The dead time input 3512 value is based on a numberof dead ticks determined based on the current command, a user setting,or both. For example, the user may input a constant value for the numberof dead ticks. As another example, the number of dead ticks may bedetermined based on the current command 2940, a voltage of power source,or both. Additionally, the dead ticks may be positive or negative andmay be in terms of either transistor, i.e., turn on the upper or lowertransistors earlier or later.

The drive enable input 3514 (DE) is configured to enable a particularlane or phase for operation or control by the PWM 242. The drive enableinput 3514 is a Boolean input or value. As illustrated in FIG. 35, thedrive enable input 3514 is determined based on a lane drive enable input3516 (e.g., [DEA]) and a shut down input 3518 ([notSD]). Each of theseinputs 3516, 3518 may include a Boolean value. The lane drive enableinput 3516 is configured to control enabling a particular lane or phaseof the motor to be active or controlled. The shut down input 3518 isconfigured to open all of the transistors 336 to deactivate the motor124. As illustrated in FIG. 35, the drive enable input 3514 is generatedas an output of an AND logic gate that receives the lane drive enableinput 3516 and the shut down input 3518. To illustrate, when the lanedrive enable input 3516 and the shut down input 3518 indicate true orlogical high, the drive enable input 3514 indicates true or high and theparticular lane is enabled for control by the PWM logic 3502.

The PWM logic 3502 includes gate drivers 3522, 3524 configured togenerate gate high and low outputs 3532, 3534 based on the high and lowactivation outputs 3506, 3508 from each lane. The gate drivers 3522,3524 may include or correspond to the gate drivers 334 of FIG. 3. Thegate high and low outputs 3532, 3534 include Boolean outputs, such astrue or false indicated by a logical 0 or a 1.

During operation, each comparison logic block 3504 generates an on timefor a corresponding lane based on the counter 3302, the comparison value3322, one or more comparison criteria (e.g., the comparison criteria3324, 3326) as described with reference to FIGS. 33 and 34. Thecomparison logic block 3504 adjusts the on time for the particular lanebased on the dead time input 3512. For example, the comparison logicblock 3504 reduces the on time for the A-lane to be high (e.g., AH) by 2ticks or counts for each dead tick of the dead time input 3512 value. Toillustrate, the upper transistor 2822 of the A-lane is turned on onecounter later (one dead tick) and is turned off one count (one deadtick) earlier than determined in FIGS. 33 and 34 so that the uppertransistor is not on at the same time as the lower transistors 2832 ofthe A-lane. Alternatively, the comparison logic block 3504 reduces theon time for the A-lane to be low (e.g., AL) by 2 dead ticks or counts.

The gate drivers 3522, 3524 generate the gate high output 3532 and thegate low output 3534 based on the high and low activation outputs 3506,3508 for each lane. The gate high output 3532 and the gate low output3534 indicate which upper and lower transistor are on or active. Asillustrated in FIG. 35, the gate drivers 3522, 3524 activate one uppertransistor (high transistor) and one lower transistor at a time. As anillustrative example, the gate drivers 3522, 3524 provide the gate highoutput 3532 and the gate low output 3534 to circuitry that generateslogical high and low signals (e.g., high and low voltages) used toactivate and deactivate the transistors 336.

FIG. 36 illustrates a method 3600 of determining rotational positionusing a dual resolver. The method 3600 may be performed by the resolver126 of FIG. 1, the resolver system 204, the demodulation system 222, thedual resolver combination system 224 of FIG. 2, the dual speed resolver312, the coarse resolver 342, the fine resolver 344 of FIG. 3, theoutput circuitry 532, the angle combination circuitry 542, the driftcorrection circuitry 544 of FIG. 5, or a combination thereof.

The method 3600 includes, at 3602, receiving coarse position signalsfrom a coarse resolver and fine position signals from a fine resolver.The coarse position signals indicative of a coarse position of a driveshaft of a motor, and the fine position signals indicative of a fineposition of the drive shaft of the motor. The coarse resolver mayinclude or correspond to 342 of FIG. 3. The coarse position signals mayinclude or correspond to the differential voltage outputs 354, thedifferential voltage signals 356, the ADC outputs 358 of FIG. 3, thesine and cosine values 1312, 1314 of FIG. 13, or a combination thereof,and the fine position signals may include or correspond to thedifferential voltage outputs 354, the differential voltage signals 356,the ADC outputs 358 of FIG. 3, the ADC output signal 1212 of FIG. 12, ora combination thereof.

The method 3600 of FIG. 36 also includes, at 3604, generating an initialposition output, based on the coarse position signals, that indicates aninitial position of the drive shaft. The initial position output mayinclude or correspond to the initial position output 2012 of FIG. 20 andthe initial position of the drive shaft may include or correspond theactual angle 2004 of FIG. 20. In some implementations, the initialposition output corresponds to an output of an initialization mode ofthe dual speed resolver 312.

The method 3600 of FIG. 36 further includes, at 3606, generating asubsequent position output, based on the fine position signals, thatindicates a subsequent position of the drive shaft. The subsequentposition output may include or correspond to the subsequent positionoutput 2014 of FIG. 20 of FIG. 20 and the subsequent position of thedrive shaft may include or correspond the actual angle 2004 of FIG. 20.In some implementations, the subsequent position output corresponds toan output of a regular operation mode (i.e., a non-initialization mode)of the dual speed resolver 312.

In some implementations, the coarse position signals and the fineposition signals are generated based on an excitation signal received bythe resolvers. In a particular implementation, the excitation signalincludes or corresponds to a dithered excitation signal, such as thedithered excitation signal 352 of FIGS. 3 and 23. Additionally oralternatively, the excitation signal (or the dithered excitation signal352) is time coordinated with current direct switching of the motor. Forexample, the excitation signal is offset from current drive switching ofthe motor 124, as described with reference to FIGS. 2 and 3. Toillustrate, the excitation signals 452 are synchronized with currentdrive switching signals associated with the motor 124 such that peakamplitudes of the excitation signals 452 are offset from current driveswitching signals.

In some implementations, the subsequent position output is generatedbased on the initial position output and the fine position signals,wherein the subsequent position output is generated independent ofsubsequent coarse position signals, as described with reference to FIGS.18 and 19.

In some implementations, generating the initial position output includesdetermining a starting position of the drive shaft based on the coarseposition signals and providing the starting position as the initialposition output based on an initialization mode input, as described withreference to FIGS. 18 and 19. For example, the logic 1802 generates thecombined estimated position 1832 based on the estimated initial position1822 (indicating the starting position determined based on resolveroutputs 354 of the coarse resolver 342) provided from the switch 1826responsive to the Boolean input 1824 (init_rslv) indicating true.

In some implementations, generating the subsequent position outputincludes differentiating the fine position signals and converting thedifferentiated fine position signals to a first domain associated withthe coarse position signals. Generating the subsequent position outputfurther includes generating an estimated subsequent position based onthe initial position output, the differentiated fine position signals,and a drift correction value and includes outputting the subsequentposition based on the estimated subsequent position, as described withreference to FIG. 18. For example, the logic 1802 differentiates thesecond domain angle estimates 1812 (the angle estimates 364 in a seconddomain based on fine position signals from the fine resolver 344) bysubtracting the previous position value 1814 from the current positionvalue 1816 of the second domain angle estimates 1812 to generate thechange in position value 1818. The logic 1802 converts the change inposition value 1818 to the converted change in position value 1820 andprovides the converted change in position value 1820 to the combiner1830. The combiner 1830 generates a subsequent combined estimatedposition 1832 based on the converted change in position value 1820, thedelayed combined estimated position 1844 (e.g., the initial positionoutput or initial combined estimated position 1832), and the driftcorrection value 1828.

In some implementations, generating the drift correction value includesconverting the initial position output to a second domain associatedwith the fine position signals and generating an error value based onthe converted initial position output, the fine position signals, and a180 degree bias offset value. Generating the drift correction valuefurther includes determining the drift correction value based on theerror value, a proportional gain, and an integral gain, as describedwith reference to FIG. 19. For example, the logic 1902 converts thedelayed combined estimated position 1844 from the 1 speed domain intothe 16 speed and subtracts the converted estimate angle 1914 and the 180degree bias offset 1918 from the second domain angle estimates 1812 togenerate the error value 1920. The logic 1902 generates the driftcorrection value 1828 based on the error value 1920, the proportionalgain 1922, and the integral gain 1924. In a particular implementation,the error value 1920 is right shifted to apply the proportional gain1922 and the integral gain 1924 corresponds to 1 or −1 based on theerror value 1920.

In some implementations, the initial position output and a plurality ofsubsequent position outputs, including the subsequent position output,are generated based on contiguous math. For example, the resolver system204 generates position outputs based on contiguous functions where inputparameters change one at a time.

In some implementations, a plurality of subsequent position outputs,including the subsequent position output, are generated independent ofmode switching or counter jumps. For example, the resolver system 204generates a plurality of subsequent position outputs corresponding tooutputs for a full rotation of the motor (0 to 360 degrees) withoutswitching modes or resetting a counter. The resolver system 204 does notneed to have a separate mode for each half or quadrant of the motor andswitch between the modes as the motor rotates from 0 to 360 degrees.

In some implementations, the method 3600 further includes transformingthe fine position signals from the fine resolver from a second domaininto a first domain associated with the coarse resolver and generatingthe subsequent position outputs based further on the transformed fineposition signals. For example, the second domain angle estimate 1812 isdifferentiated to produce the change in position value 1818; the changein position value 1818 is transformed from the second domain to thefirst domain by right shifting the change in position value 1818 by fourto generate the converted change in position value 1820 (transformedfine position signals), as described with reference to FIG. 18.

In some implementations, the method 3600 further includes transformingthe initial position output into a second domain associated with thefine resolver and generating the subsequent position output basedfurther on the transformed initial position output. For example, thedelayed combined estimated position 1844 is transformed from the firstdomain to the second domain by left shifting the delayed combinedestimated position 1844 by four to generate the converted estimate angle1914 (transformed initial position output), as described with referenceto FIG. 19.

FIG. 37 illustrates a method 3700 of pulse-width modulation. The method3700 may be performed by the inverter 112 of FIG. 1, the PWM 242 of FIG.2, the PWM logic 3502 of FIG. 35, or a combination thereof. The method3700 includes, at 3702, receiving a comparison value and a comparisoncriterion. The comparison value may include or correspond to comparisonvalue 3322 of FIG. 33 or the comparison value 3422 of FIG. 34, and thecomparison criterion may include or correspond to a comparison criterionof the comparison criteria 3324, 3326, 3424, or 3426 of FIGS. 33 and 34.In some implementations, the comparison value and the comparisoncriterion are included in a set point signal, such as the set pointsignals 3312-3320 and 3412-3420 of FIGS. 33 and 34.

The method 3700 of FIG. 37 also includes, at 3704, comparing thecomparison value to a counter value based on the comparison criterion.The counter value may include or correspond to a counter value or of thecounter signal of FIG. 33. The counter value may be received from acounter, such as the counter 3302 of FIG. 33.

The method 3700 of FIG. 37 further includes, at 3706, in response to thecomparison value satisfying the comparison criterion with respect to thecounter value, sending a control signal to a gate of a first transistorto generate a pulse edge of a pulse of a pulse-width modulated signal.The control signal may include or correspond to the one of the high andlow activation outputs 3506, 3508 or the gate high and low outputs 3532,3534 of FIG. 35. The gate of the first transistor may include orcorresponds to a gate of any of the transistors 336 of FIG. 3 or thetransistors 2822-2826 or 2832-2836 of FIG. 28. The pulse edge of thepulse-width modulated signal 378 may include or correspond to a firstedge or a second edge of the pulse-width modulated signal 378 of FIG. 3.

In some implementations, the method 3700 of FIG. 37 includes receiving aset point signal indicating the comparison value, the comparisoncriterion, and a second comparison criterion and comparing thecomparison value to a second counter value based on the secondcomparison criterion. The method 3700 further includes, in response tothe comparison value satisfying the second comparison criterion withrespect to the second counter value, sending a second control signal tothe gate of the first transistor to generate a second pulse edge of thepulse of the pulse-width modulated signal, as described with referenceto FIGS. 33 and 34. For example, the PWM 242 receives the first setpoint signal 3312 and generates the second pulse edge of the first pulsebased on comparing the comparison value 3322 to a value of the counter3302 based on the second comparison criterion 3326.

In some implementations, the method 3700 of FIG. 37 includes determiningthe comparison value based on one or more first bits of the set pointsignal and determining the comparison criterion and the secondcomparison criterion based on one or more second bits of the set pointsignal, as described with reference to FIGS. 33 and 34. For example, thePWM 242 receives the first set point signal 3312 and generates thecomparison value 3322 by right shifting a value of the first set pointsignal 3312 indicated by the first set point signal 3312 (e.g.,indicated by all of the bits of the first set point signal) andgenerates the comparison values by determining a value of LSBs of thefirst set point signal 3312 and retrieving the comparison criteria 3324,3326 based on the value of the LSBs.

In some implementations, determining the comparison value includes rightshifting the set point signal by one or more bits, as described withreference to FIGS. 33 and 34. In some implementations, the one or moresecond bits of the set point signal correspond to one or more leastsignificant bits of the set point signal, as described with reference toFIGS. 33 and 34. In some implementations, the method 3700 of FIG. 37includes selecting the comparison criterion from a table based on theone or more least significant bits of the set point signal, as describedwith reference to FIGS. 33 and 34.

In some implementations, the method 3700 of FIG. 37 includes receiving asecond set point signal indicating a second comparison value and asecond comparison criterion and comparing a second comparison value tothe second counter value based on the second comparison criterion. Themethod 3700 further includes, in response to the second comparison valuesatisfying the second comparison criterion with respect to the secondcounter value, sending a second control signal to a gate of a secondtransistor to generate a pulse edge of a second pulse of the pulse-widthmodulated signal, as described with reference to FIGS. 33 and 34. Forexample, the PWM 242 receives the second set point signal 3314 andgenerates the first pulse edge of the second pulse based on comparingthe comparison value 3322 to a value of the counter 3302 based on thefirst comparison criterion 3324 of the second set point signal 3314.

In some implementations, the method 3700 of FIG. 37 includes receiving asecond set point signal indicating a second comparison value and asecond comparison criterion and comparing a second comparison value tothe second counter value based on the second comparison criterion. Themethod 3700 further includes, in response to the second comparison valuesatisfying the second comparison criterion with respect to the secondcounter value, sending a second control signal to a gate of a secondtransistor to generate a second pulse edge of the pulse of thepulse-width modulated signal, as described with reference to FIGS. 33and 34. For example, the PWM 242 receives the first set point signal3312 and generates the second pulse edge of the first pulse based oncomparing the comparison value 3422 to a value of the counter 3302 basedon the second comparison criterion 3426.

FIG. 38 illustrates a method 3800 for feedback control. The method 3800may be performed by the control system 205, the feedback control system232 of FIG. 2, the logic 3002 of FIG. 30, the logic 3102 of FIG. 31, thelogic 3202 of FIG. 32, or a combination thereof. The method 3800includes, at 3802, receiving a position command for a controlledcomponent and a speed command for the controlled component. Thecontrolled component may include or correspond to the motors 124, adrive shaft of the motors 124, the gimbals 122, the sensors 128 of FIG.1, or a combination thereof. The position command may include orcorrespond to position command 2922 of FIGS. 29-32. The speed commandmay include or correspond to speed command 2932 of FIGS. 29-32. In someimplementations, the position command indicates an angle of the motorand the speed command indicates an RPM of the motor.

The method 3800 of FIG. 38 also includes, at 3804, generating a ratelimited position command based on the speed command and the positioncommand. The rate limited position command may include or correspond tothe rate limited position command 3034 of FIGS. 30-32. In someimplementations, the method 3800 of FIG. 38 includes converting thecurrent command to a duty cycle value and controlling a motor based onthe duty cycle value, as described with reference to FIGS. 3 and 33-35.

The method 3800 of FIG. 38 includes, at 3806, receiving positionfeedback indicating a position of the controlled component. The positionfeedback may include or correspond to the position feedback 2924 ofFIGS. 29-32. In some implementations, the position feedback is generatedbased on resolver outputs. For example, the position feedback may begenerated by a resolver practicing the method 3600 of FIG. 36, by ademodulation circuit practicing the method 3900 of FIG. 39, or both.

The method 3800 of FIG. 38 further includes, at 3808, applying a controlgain to an error signal to generate an adjusted error signal. The errorsignal is based on the position feedback and the rate limited positioncommand. The control gain may include or correspond to the proportionalgain 2926, the integral gain 2928 of FIGS. 29-32, or both. The errorsignal may include or correspond to the error signal 3012, and theadjusted error signal may include or correspond to the adjusted errorsignal 3014 of FIGS. 30-32.

The method 3800 of FIG. 38 further includes, at 3810, outputting acurrent command based on the adjusted error signal. The current commandmay include or correspond to the current command 2940 of FIGS. 29-32. Insome implementations, the current command is used to control a motor andindicates a torque of the motor, as described with reference to FIG. 3.For example, the current command 2940 is provided to the current tracker330, and the current tracker 330 generates the duty cycle value 374based on the current command 2940, as described with reference to FIG.3. The PWM 242 receives a set point signal indicating the duty cyclevalue 374 of the duty cycle setting 376 and supplies the AC power signal380 to the motor 120 based on the set point signal.

In some implementations, generating the rate limited position commandbased on the speed command and the position command includes generatinga threshold (e.g., a threshold value) based on the speed command andreducing a value of the position command to value of the speed commandto generate the limited position command, as described with reference toFIGS. 30-32.

In some implementations, the method 3800 of FIG. 38 includes generatingthe error signal based on the position feedback and the rate limitedposition command. In some such implementations, generating the errorsignal includes subtracting the position indicated by the positionfeedback signal from the position indicated by the rate limited positioncommand, as described with reference to FIGS. 30-32.

In some implementations, applying the control gain to the error signalto generate the adjusted error signal includes applying a proportionalgain to the error signal and applying an integral gain to the errorsignal. Additionally or alternatively, applying the control gain to theerror signal to generate the adjusted error signal includes multiplyingthe error signal by a proportional gain to generate a first product,multiplying an integral of the error signal by an integral gain togenerate a second product, and adding the first product and the secondproduct to generate the adjusted error signal, as described withreference to FIGS. 30-32.

In some implementations, the method 3800 of FIG. 38 includes dampeningthe adjusted error signal based on speed feedback, such as the speedfeedback 2936 of FIGS. 29-32. In some such implementations, the currentcommand is generated based on the damped error signal, such as thedamped error signal 3138 of FIG. 31 and generating the damped errorsignal includes applying a second control gain to the adjusted errorsignal, as described with reference to FIGS. 30-32.

In a particular implementation, applying the second control gain to theadjusted error signal includes differentiating the speed feedback togenerate differentiated speed feedback, multiplying the differentiatedspeed feedback by the second control gain to generate a damping value,and subtracting the damping value from the adjusted error signal, wheresubtracting the damping value from the adjusted error signal generatesthe damped error signal, and where the current command is generatedbased on the damped error signal, as described with reference to FIGS.30-32. For example, the speed feedback 2936 is differentiated andmultiplied by the rpm gain 2938 to generate the damping value 3016. Thedamping value 3016 is subtracted from the adjusted error signal 3014 togenerate the damped error signal 3138; and the damped error signal 3138is limited by minimum and maximum values and the limited damped errorsignal 3138 integrated to generate the current command 2940, asdescribed with reference to FIGS. 30-32.

FIG. 39 illustrates a method 3900 of demodulating resolver outputs. Themethod 3900 may be performed by the resolver system 204, thedemodulation system 222, the dual resolver combination system 224, thedemodulation circuitry 514, 524 of FIG. 5, one or more circuits orfirmware configured to perform operations of the logic of FIGS. 7-11,14, 16, 18, and 19, or a combination thereof. The method 3900 includes,at 3902, receiving, from an analog-to-digital converter (ADC), aplurality of ADC outputs. The plurality of ADC outputs are generatedbased on the resolver outputs. The ADC may include or correspond to oneof the ADCs 318, 320 of FIG. 3. The resolver may include or correspondto the resolvers 126 of FIG. 1, the dual speed resolver 312, the coarseresolver 342, the fine resolver 344 of FIG. 3, or a combination thereof.The ADC outputs may include or correspond to the ADC outputs 358 or theconditioned voltage values 360 of FIG. 3.

The method 3900 of FIG. 39 also includes, at 3904, rectifying theplurality of ADC outputs based on a square wave. In someimplementations, the square wave includes a square wave having anamplitude of 1 and −1. The square wave may switch between amplitudesbased on a counter value, such as the flip input 1022 which isdetermined based on the counter input 852, as described with referenceto FIGS. 10-12.

The method 3900 of FIG. 39 includes, at 3906, determining, based on therectified plurality of ADC outputs, demodulated amplitude valuescorresponding to the resolver outputs. The demodulated amplitude valuesmay include or correspond to the demodulated amplitude values 362 ofFIG. 3. In some implementations, the demodulated amplitude values may begenerated by recursive median filtering, as described with reference toFIGS. 9 and 16.

The method 3900 of FIG. 39 further includes, at 3908, generatingposition outputs based on the demodulated amplitude values. The positionoutputs may include or correspond to the angle estimates 364 or theposition estimate 366 of FIG. 3. In some implementations, the positionoutputs indicate a position (angle) of the drive shaft of the motor 124,as described with reference to FIG. 3. In some implementations, thedemodulated amplitude values include sign information (e.g., are signedintegers) and the angle estimates 364, the position estimates 366, orboth are calculated based on the function atan 2. Additionally oralternatively, the position outputs may be generated based on sharedsine table circuitry.

In some implementations, generating the demodulated amplitude valuesincludes generating accumulated values based on the rectified pluralityof ADC outputs, computing a median value of the last n number ofaccumulated values, and outputting the median value as a particulardemodulated amplitude value of the demodulated amplitude values, asdescribed with reference to FIG. 16.

In some implementations, rectifying the plurality of ADC outputsincludes multiplying the plurality of ADC outputs based on a squarewave. In a particular implementation, the square wave has an amplitudeof 1 for counter values below a particular counter value, such as amedian counter value, and has an amplitude of −1 for counter valuesabove the particular counter value, as described with reference to FIGS.10-12.

In some implementations, multiplying the plurality of ADC output basedon the square wave includes changing a sign of a particular ADC outputof the plurality of ADC outputs to generated an inverted ADC output,selecting the particular ADC output or the inverted ADC output based ona flip input, and outputting the selected ADC output as a particularrectified ADC output, as described with reference to FIG. 11.

In some implementations, the method 3900 of FIG. 39 includes masking asubset of ADC outputs that align with current drive switching of a motorassociated with the resolver to generate a filtered set of ADC outputs.The method 3900 further includes determining the demodulated amplitudevalue based on the filtered set of ADC outputs in the data windows 1514,as described with reference to FIGS. 14 and 15.

In some implementations, the masked set of ADC output comprises ADCoutputs that occur near peak amplitudes of a first harmonic of anexcitation signal associated with the resolver, as described withreference to FIGS. 14 and 15. Additionally, each demodulated amplitudevalue of the demodulated amplitude values corresponds to a cycle of thefirst harmonic of the excitation signal, as described with reference toFIGS. 14-16.

In some implementations, the method 3900 of FIG. 39 includes recursivelyfiltering the plurality of ADC outputs to generate median ADC outputs,as described with reference to FIGS. 8 and 9. In such implementations,rectifying the plurality of ADC outputs includes rectifying the medianADC outputs.

In some implementations, the resolver receives a dithered excitationsignal that has zero mean dither and generates the resolver outputsbased on the dithered excitation signal. In such implementations, theADC oversamples the resolver outputs to generate the ADC output.

FIG. 40 illustrates a method 4000 of generating an excitation signal fora sensor device. The method 4000 may be performed by the excitationsignal generation system 202, the dither generator 212, the coordinationsystem 214 of FIG. 2, the logic 2602 of FIG. 26, the resolver drivercircuit 2702 of FIG. 27, or a combination thereof. The method 4000includes, at 4002, generating a high order even harmonic of a baseexcitation signal. The high order even harmonic may include orcorrespond to the high order even harmonic signal 2666 of FIG. 26. Thebase excitation signal may include or correspond to the base excitationsignal 452 of FIG. 23.

The method 4000 of FIG. 40 also includes, at 4004, generating a ditheredexcitation signal based on combining the high order even harmonic andthe base excitation signal. The dithered excitation signal may includeor correspond to the dithered excitation signal 352 of FIGS. 3 and 23.

The method 4000 of FIG. 40 further includes, at 4006, outputting thedithered excitation signal to the sensor device. In someimplementations, the sensor device comprises a resolver. In a particularimplementation, the resolver comprises a dual resolver or dual speedresolver, such as the dual speed resolver 312 of FIG. 3. In otherimplementations, the sensor device may include or correspond to asynchro, active transducers (e.g., a linear variable differentialtransformer), or other active sensors. Additionally, the sensor may beincluded in the inertial measurement unit 102 of FIG. 1.

In some implementations, the dithered excitation signal includes orcorresponds to a digital signal and includes digital samples. In suchimplementations, a DAC, such as the DAC 310, converts the ditheredexcitation signal into an analog signal and provides the ditheredexcitation signal to the sensor device.

In some implementations, the base excitation signal corresponds to asine wave signal and the dithered excitation signal has zero meandeviation from the sine wave signal. For example, a 16th order harmonicsignal includes 16 sine waves for each sine wave of the base excitationsignal. Thus, a combined signal of the high order even harmonic and thebase excitation signal produces the dithered excitation signal 352 whereduring each sine wave of the base excitation signal 452, an averageamplitude of the dithered excitation signal 352 does not deviate from anaverage amplitude of the base excitation signal 452, as illustrated inFIG. 23.

In some implementations, the method 4000 further includes generating thebase excitation signal. For example, the excitation signal generatorsystem 202 of FIG. 2 generates the base excitation signal 452 andprovides the base excitation signal 452 to the combiners 2624, 2626, asdescribed with reference to FIG. 26. In a particular implementation, theexcitation signal is generated based on a sine function and an amplitudesetting. For example, the excitation signal generation system 202 (e.g.,the first logic chain 2612 thereof) generates the base excitation signal452 by retrieving a value from a sine look-up table and multiplying thevalue by the amplitude setting. In some such implementations, thelook-up table is shared with one or more other components, the look-uptable is accessed based on Round-Robin scheduling.

In some implementations, the method 4000 further includes generating thehigh order even harmonic based on a 16^(th) harmonic of the sinefunction and a second amplitude setting. For example, the second logicchain 2614 of the excitation signal generation system 202 generates thehigh order even harmonic signal 2666 based on the harmonic of the sinefunction and the amplitude setting and provides the high order evenharmonic signal 2666 to the combiners 2624, 2626, as described withreference to FIG. 26.

In some implementations, the method 4000 further includes generating alow order odd harmonic signal and combining the low order odd harmonic,the high order even harmonic, and the base excitation signal to generatethe dithered excitation signal. In other implementations, the combineris configured to combine the high order even harmonic and the baseexcitation signal to generate the dithered excitation signal.

In some implementations, the method 4000 further includes timecoordinating the dithered excitation signal with current switching of amotor. In a particular implementation, time coordinating the ditheredexcitation signal with current switching of the motor includesoffsetting current switching from peaks and valleys of the baseexcitation signal, as described with reference to FIGS. 27, 28, and33-35. Additionally or alternatively, time coordinating the ditheredexcitation signal with current switching of the motor includesperforming an even number of current switches during a wave of the baseexcitation signal, as described with reference to FIG. 33. In aparticular implementation, the number of current switched occurringduring a wave has an equal number of current switch activations andcurrent switch deactivations.

In some implementations, the method 4000 further includes timecoordinating the dithered excitation signal with current switching of amotor. In a particular implementation, time coordinating the ditheredexcitation signal with current switching of the motor includesoffsetting current switching from peaks and valleys of the baseexcitation signal, as described with reference to FIG. 33. Additionallyor alternatively, the dithered excitation signal is synched with asampling rate of outputs of the resolver.

In some implementations, a motor is coupled to the sensor and thedithered excitation signal is offset the current drive switching of themotor. In some such implementations, the method 4000 further includestime coordinating a first clock, used to generate the ditheredexcitation signal, with a second clock of a PWM, configured to controlthe motor, such that current drive switching of the motor is offset fromthe dithered excitation signal. For example, the counters 842 of FIG. 8and 3302 of FIG. 33 may be synchronized.

In some implementations, an ADC is coupled to the resolver. In some suchimplementations, the method 4000 further includes receiving, by the ADC,output signals from the resolver and oversampling the output signals. Ina particular implementation, the first clock is synchronized with asecond clock of the ADC. In other implementations, the ADC oversamplesthe output signals based on the first clock, such as the counter 842 ofFIG. 8.

Referring to FIG. 41, a block diagram 4100 of an illustrative embodimentof an aircraft 4102. As shown in FIG. 41 the aircraft 4102 (e.g., aspaceship, a satellite, or a space station) includes a frame 4118, aninterior 4122, and a plurality of systems 4120. The systems 4120 mayinclude one or more of a propulsion system 4124, an electrical system4126, an environmental system 4128, a hydraulic system 4130, and theinertial measurement unit 102. In other implementations, the aircraft4102 may include additionally systems or fewer systems than the systems4120 illustrated in FIG. 41.

The inertial measurement unit 102 includes the inverter 112 and thegimbal device 114 of FIG. 1. The inverter 112 may include the PWM 242 ofFIG. 2, the current tracker 330, the gate drivers 334, the transistors336 of FIG. 3, the motor driver circuit 2802 of FIG. 28, or acombination thereof. The inverter 112 may include circuitry and firmwareconfigured to perform the operations described in FIG. 33, theoperations described in FIG. 34, the operations of the PWM logic 3502 ofFIG. 35, or a combination thereof.

The gimbal device 114 may include the gimbals 122, the motors 124, theresolvers 126, the sensors 128 of FIG. 1, the excitation signalgeneration system 202, the resolver system 204, the control system 206,the demodulation system 222, the dual resolver combination system 224,the feedback control system 232 of FIG. 2, the DAC 310, the dual speedresolver 312, the differential voltage sensors 314, 316, the ADCs 318,320, the voltage conditioning circuitry 322, 324, the angle estimationcircuitry 326 of FIG. 3, the demodulation circuitry 514, 524, the anglecalculation circuitry 516, 526, the angle combination circuitry 542, andthe drift correction circuitry 544 of FIG. 5, or a combination thereof.

Additionally or alternatively, the gimbal device 114 may includecircuitry and firmware configured to perform operations of the logic 602of FIG. 6, the logic 702 of FIG. 7, the logic 802 of FIG. 8, the RMVAlogic 812 of FIG. 9, the demodulation logic 826 of FIG. 10, theaccumulator logic 1014 of FIG. 11, the logic 1102 of FIG. 11, the masklogic 1018 of FIG. 14, the output logic 1026 of FIG. 16, the logic 1802of FIG. 18, the logic 1902 of FIG. 19, the logic 2602 of FIG. 26, thelogic 3002 of FIG. 30, the logic 3102 of FIG. 31, the logic 3202 of FIG.32, or a combination thereof. The inertial measurement unit 102 may beincluded within in the aircraft 4102 or coupled to an exterior surfaceof the aircraft 4102.

The inertial measurement unit 102 is configured to enable inertialnavigation of the aircraft 4102 as described above with reference toFIGS. 1-40. For example, the inertial measurement unit 102 (e.g., one ormore components thereof) is configured to perform one or more operationsof one or more of the method 3600-400. As another example, the inertialmeasurement unit 102 is be configured to execute computer-executableinstructions (e.g., a program of one or more instructions) stored in amemory. The instructions, when executed, cause the inertial measurementunit 102 to perform one or more operations of one or more of the method3600-4000. For example, the processor receives the coarse positionsignals from the coarse resolver and the fine position signals from thefine resolver, generates the initial position output based on the coarseposition signals, and generates the subsequent position output based onthe fine position signals, as described with reference to FIG. 36.

The illustrations of the examples described herein are intended toprovide a general understanding of the structure of the variousimplementations. The illustrations are not intended to serve as acomplete description of all of the elements and features of apparatusand systems that utilize the structures or methods described herein.Many other implementations may be apparent to those of skill in the artupon reviewing the disclosure. Other implementations may be utilized andderived from the disclosure, such that structural and logicalsubstitutions and changes may be made without departing from the scopeof the disclosure. For example, method operations may be performed in adifferent order than shown in the figures or one or more methodoperations may be omitted. Accordingly, the disclosure and the figuresare to be regarded as illustrative rather than restrictive.

Moreover, although specific examples have been illustrated and describedherein, it should be appreciated that any subsequent arrangementdesigned to achieve the same or similar results may be substituted forthe specific implementations shown. This disclosure is intended to coverany and all subsequent adaptations or variations of variousimplementations. Combinations of the above implementations, and otherimplementations not specifically described herein, will be apparent tothose of skill in the art upon reviewing the description.

The Abstract of the Disclosure is submitted with the understanding thatit will not be used to interpret or limit the scope or meaning of theclaims. In addition, in the foregoing Detailed Description, variousfeatures may be grouped together or described in a single implementationfor the purpose of streamlining the disclosure. Examples described aboveillustrate but do not limit the disclosure. It should also be understoodthat numerous modifications and variations are possible in accordancewith the principles of the present disclosure. As the following claimsreflect, the claimed subject matter may be directed to less than all ofthe features of any of the disclosed examples. Accordingly, the scope ofthe disclosure is defined by the following claims and their equivalents.

What is claimed is:
 1. Demodulation circuitry comprising: an inputterminal configured to be coupled to an analog-to-digital converter(ADC) and configured to receive a plurality of ADC outputs, wherein theplurality of ADC outputs are generated based on resolver outputs; arectifier configured to rectify the plurality of ADC outputs, whereinrectifying the plurality of ADC outputs preserves a phase of theplurality of ADC outputs; amplitude determination circuitry configuredto determine, based on the rectified plurality of ADC outputs,demodulated amplitude values corresponding to the resolver outputs; andangle computation circuitry configured to determine estimated angleoutputs based on the demodulated amplitude values and configured togenerate position outputs based on the estimated angle outputs.
 2. Thedemodulation circuitry of claim 1, wherein the rectifier is configuredto rectify the plurality of ADC outputs by multiplying the plurality ofADC outputs by a square wave.
 3. The demodulation circuitry of claim 1,further comprising a second input terminal configured to be coupled to asecond ADC and configured to receive a second plurality of ADC outputs,wherein the second plurality of ADC outputs are generated based onoutputs of a second resolver of a dual speed resolver.
 4. Thedemodulation circuitry of claim 1, wherein the demodulated amplitudevalues include sign information, and wherein the angle computationcircuitry is configured to compute the estimated angle outputs using anatan 2 function.
 5. The demodulation circuitry of claim 1, wherein theangle computation circuitry is configured to generate the estimatedangle outputs based on shared sine table circuitry, and wherein theshared sine table circuitry is shared with excitation signal generationcircuitry via Round-Robin scheduling.
 6. The demodulation circuitry ofclaim 1, wherein the amplitude determination circuitry comprises anaccumulator configured to accumulate multiple rectified ADC outputs ofthe rectified plurality of ADC outputs to generate accumulated outputs,and wherein the demodulated amplitude values are generated based on theaccumulated outputs.
 7. The demodulation circuitry of claim 6, furthercomprising a counter configured to generate a counter value, wherein theaccumulator is configured to output a particular accumulated output as aparticular demodulated amplitude value responsive to the counter havinga particular counter value.
 8. The demodulation circuitry of claim 1,further comprising voltage conditioning circuitry configured to adjust avoltage of the plurality of ADC outputs to account for a voltage bias ofthe demodulation circuitry, the voltage conditioning circuitrycomprising a low pass filter configured to determine the voltage bias.9. A method of demodulating resolver outputs, the method comprising:receiving, from an analog-to-digital converter (ADC), a plurality of ADCoutputs, wherein the plurality of ADC outputs are generated based on theresolver outputs; rectifying the plurality of ADC outputs, whereinrectifying the plurality of ADC outputs preserves a phase of theplurality of ADC outputs; determining, based on the rectified pluralityof ADC outputs, demodulated amplitude values corresponding to theresolver outputs; determining estimated angle outputs based on thedemodulated amplitude values; and generating position outputs based onthe estimated angle outputs.
 10. The method of claim 9, whereindetermining the demodulated amplitude values comprises: generatingaccumulated values based on the rectified plurality of ADC outputs;computing a median value of a last n number of accumulated values,wherein n is an integer greater than one; and outputting the medianvalue as a particular demodulated amplitude value of the demodulatedamplitude values.
 11. The method of claim 9, wherein rectifying theplurality of ADC outputs comprises multiplying a signal of the pluralityof ADC outputs and a square wave, wherein the square wave has anamplitude of 1 for counter values below a particular counter value andan amplitude of −1 for counter values above the particular countervalue, and wherein each ADC output of the plurality of ADC outputscorresponds to a counter value.
 12. The method of claim 11, whereinmultiplying the signal of the plurality of ADC outputs based on thesquare wave includes: changing a sign of a particular ADC output of theplurality of ADC outputs to generate an inverted ADC output; selectingthe particular ADC output or the inverted ADC output based on a flipinput, the flip input generated based on the counter values; andoutputting the selected ADC output as a particular rectified ADC outputof the rectified plurality of ADC outputs.
 13. The method of claim 9,further comprising: masking a subset of ADC outputs of the plurality ofADC outputs that align with current drive switching of a motorassociated with a resolver to generate a filtered set of ADC outputs;and determining the demodulated amplitude values based on the filteredset of ADC outputs.
 14. The method of claim 13, wherein the filtered setof ADC outputs comprises ADC outputs that occur near peak amplitudes ofa first harmonic of an excitation signal associated with the resolver,and wherein each demodulated amplitude value of the demodulatedamplitude values corresponds to a cycle of the first harmonic of theexcitation signal.
 15. The method of claim 9, further comprisingrecursively filtering the plurality of ADC outputs to generate medianADC outputs, wherein rectifying the plurality of ADC outputs includesrectifying the median ADC outputs.
 16. A system comprising: a resolver;an analog-to-digital converter (ADC) coupled to the resolver; anddemodulation circuitry coupled to the ADC and configured to generatedemodulated resolver outputs, the demodulation circuitry comprising: aninput terminal coupled to the ADC and configured to receive a pluralityof ADC outputs, wherein the plurality of ADC outputs are generated basedon resolver outputs; a rectifier configured to rectify the plurality ofADC outputs, wherein rectifying the plurality of ADC outputs preserves aphase of the plurality of ADC outputs; amplitude determination circuitryconfigured to determine, based on the rectified plurality of ADCoutputs, demodulated amplitude values corresponding to the resolveroutputs; and angle computation circuitry configured to determineestimated angle outputs based on the demodulated amplitude values andconfigured to generate position outputs based on the estimated angleoutputs.
 17. The system of claim 16, wherein the resolver is configuredto receive a dithered excitation signal that has zero mean dither and isconfigured to generate the resolver outputs based on the ditheredexcitation signal, and wherein the ADC is configured to oversample theresolver outputs to generate the plurality of ADC outputs.
 18. Thesystem of claim 16, wherein the resolver, the ADC, and the demodulationcircuitry are included in a gimbaled inertial measurement unit.
 19. Thesystem of claim 18, wherein the gimbaled inertial measurement unitfurther comprises a motor, the motor configured to position a gimbal ofthe gimbaled inertial measurement unit, and wherein the resolver isconfigured to generate the resolver outputs indicative of a position ofa drive shaft of the motor.
 20. The system of claim 19, wherein thegimbaled inertial measurement unit further comprises feedback controlcircuitry coupled to the demodulation circuitry, wherein the feedbackcontrol circuitry is configured to receive the position outputs and tocontrol a pulse-width modulation control circuit based on the positionoutputs, and wherein the pulse-width modulation control circuit isconfigured to control the motor.